Sinusoidal drive system and method for phototherapy

ABSTRACT

The LEDs in a phototherapy LED pad are controlled so that the intensity of the light varies in accordance with a sinusoidal function, thereby eliminating the harmonics that are generated when the LEDs are pulsed digitally, in accordance with a square-wave function. This is accomplished analogically by using a sinusoidal wave to control the gate of a MOSFET connected in series with the LEDs or by using a digital-to-analog converter to control the gate of the MOSFET with a stair step function representative of the values of a sinusoidal function at predetermined intervals. Alternatively, pulse-width modulation is used to control the gate of the MOSFET in such a way that the average current through the LEDs simulates a sinusoidal function. In additional to using a simple sine wave function, the LED current may also be controlled in accordance with “chords” containing multiple sine waves of different frequencies.

SCOPE OF INVENTION

This invention relates to biotechnology for medical applications,including photobiomodulation, phototherapy, and bioresonance.

BACKGROUND OF INVENTION Introduction

Biophotonics is the biomedical field relating to the electronic controlof photons, i.e. light, and its interaction with living cells andtissue. Biophotonics includes surgery, imaging, biometrics, diseasedetection, and phototherapy. Phototherapy is the controlled applicationof light photons, typically infrared, visible and ultraviolet light formedically therapeutic purposes including combating injury, disease, andimmune system distress. More specifically, phototherapy involvessubjecting cells and tissue undergoing treatment to a stream of photonsof specific wavelengths of light either continuously or in repeateddiscontinuous pulses to control the energy transfer and absorptionbehavior of living cells and tissue.

History of Pulsed Phototherapy Technology

For more than a century, doctors, researchers, and amateurexperimentalists have dabbled with the response of living cells andtissue to non-ionizing energy, including ultraviolet and visible light,infrared light and heat, microwaves, radio waves, alternating current(specifically microcurrents), ultrasound and sound. In many cases, theenergy source is modulated with oscillations or pulses, reportedlyresulting in “biomodulation” effects that are different from the effectsresulting from the steady application of energy. Even the famousscientist and father of alternating current Nicholas Tesla was known tohave subjected himself to high-frequency modulated electrical shocks or“lightning strikes” in theatrical public demonstrations to showcase thesupposed benefits of AC technology and oscillatory energy.Unfortunately, despite all the interest and activity, rather thanproducing a systematic comprehensive knowledge of the cellularinteractions with constant and oscillatory directed energy, theconsequence of these sensationalized and poorly controlled experimentshas produced a confusing, and even self-contradictory, mix of science,pseudo-science, mysticism, and religion. Promulgating these conflictingand sometimes extraordinary claims, today's publications, literature,and web sites range from hard science and biotechnology research toholistic medicine and spiritualism, and often represent sensationalpseudo-science (devoid of technical evidence) purely for the purpose ofenticing clients and promoting product sales.

Topically, while the greatest interest in directed-energy therapy todayis focused on low-level pulsed light for healing (i.e. phototherapy),the earliest studies concerning the influence of oscillatory energy onthe process of healing in animal and human tissue did not utilize light,but instead involved stimulating tissue with sinusoidal electricalmicrocurrents. Performed by the acupuncturist Dr. Paul Nogier in the mid1950s, this poorly-documented empirically-based work concluded thatcertain frequencies stimulate healing faster than others and manifesttissue-specificity. The studies were performed in the audio frequencyrange from zero (DC) to 20 kHz.

Absent clear documentation of the treatment conditions and the apparatusemployed, to our knowledge, exact scientific reproduction of Nogier'sexperiments and verification of his results have not occurred and noscientific technical reports appear in the refereed publishedliterature. So rather than constituting a specific method to curedisease or combat pain, Nogier's reported observations have served as aroadmap, i.e. a set of guiding principles, in the subsequent explorationand development of the field, including the following premises:

-   -   In human patients, the healing of injured or diseased tissue and        a patient's perceived pain varies with the oscillating frequency        of electrical stimulation (especially 292 Hz or “D” in the        musical scale)    -   Specific frequencies in the audio range of 20 kHz and below,        appear to stimulate different tissue and organs more than        others, i.e. tissue specificity is frequency dependent    -   Doubling a given frequency appears to behave similar to the        original frequency in tissue specificity, in effect, and in        efficacy.        It is curious to note in the last bullet point, that        even-multiples of a frequency behave similarly, implies harmonic        behavior in cellular biology and physiological processes. Such        harmonic behavior is analogous to the design of a piano and its        keyboard, where doubling or halving a frequency is musically        equivalent to the same note one octave, i.e. eight whole tones,        higher or lower than the original. Also, the reported benefit of        “even” harmonics is consistent with mathematical analysis of        physical systems showing even-harmonics couple energy more        efficiently, and behave more predictably than circuits or        systems exhibiting odd harmonics.

While Nogier's observations have become a serious research topic in themedical research community (especially in its applicability tophototherapy), they also have fueled fanatical claims promotinghighly-dubious metaphysical and religious principles that life comprisesa single pure frequency, that anything that disturbs that frequencyrepresents disease or injury, and that eliminating or cancelling thesebad frequencies somehow will restore health. Even though suchincredulous claims for maintaining health have been debunkedscientifically, proponents of this theory continue to offer for profitproducts or services for “enhancing” a person's healthy frequency usingso-called “bioresonance” for better health and longer life.

In the context of this application, any discussion of bioresonanceherein does not refer to this metaphysical interpretation of the wordbut instead refers to well defined biochemical processes in cells and intissue resulting from photobiomodulation. In fact, scientificmeasurements reveal that not one, but many dozens of frequenciessimultaneously coexist in a human body. These measured frequencies—somerandom, some fixed frequency, and some time-varying, exist mostly in theaudio spectrum, i.e. below 20 kHz. These naturally occurring frequenciesinclude ECG signals controlling heart function, EEG signals in the braincontrolling thought, visual signals carried by the optic nerve,time-varying muscle stimulation in the peripheral muscles, peristalticmuscle contractions in the intestines and uterus, nerve impulses fromtactile sensations carried by the central nervous system and spinalcord, and more. Similar signals are observed in humans, other mammalsand in birds. So clearly there is no one frequency that uniformlydescribes a healthy condition for life.

Starting in the late 1960s, medical interest turned from microcurrentsto phototherapy, as pioneered by the Russians and the Czechs and laterin the 1980s by NASA-sponsored research in the United States. In thecourse of researching phototherapy, also known as low-level lighttherapy (LLLP), the same question of modulating frequency arose,comparing pulsed light to continuous irradiation for phototherapytreatment. The efforts primarily focused on red and infrared lightpulsed at frequencies in the audio range, i.e. below 20 kHz.

Numerous studies and clinical trials have since compared various pulsedinfrared laser methods to continuous wave treatments for phototherapy.In the journal paper “Effect of Pulsing in Low-Level Light Therapy”published in Lasers Surg. Med. August 2010, volume 42(6), pp. 450-466,the authors and medical doctors Hashmi et al. from Massachusetts GeneralHospital, Harvard Medical School, and other hospitals, criticallyreviewed nine direct comparative trials of pulsed wave (PW) andcontinuous wave (CW) tests. Of these trials, six studies showed pulsedtreatments outperformed continuous illumination, and only in two casesdid the continuous wave treatment outperform light pulsing. In thesepublished works, however, no agreement or consensus was reached definingthe optimum pulse conditions for therapeutic efficacy.

One such study showing that pulsed-light phototherapy outperformscontinuous light, published in Laser Medical Science, 10 Sep. 2011,entitled “Comparison of the Effects of Pulsed and Continuous Wave Lighton Axonal Regeneration in a Rat Model of Spinal Cord Injury,” by X. Wuet al. addresses the subject of nerve repair. Excerpts include anintroduction stating: “Light therapy (LT) has been investigated as aviable treatment for injuries and diseases of the central nervous systemin both animal trials and in clinical trials. Based on in vivo studies,LT has beneficial effects on the treatment of spinal cord injury (SCI),traumatic brain injury, stroke, and neurodegenerative diseases.”

The study then concentrated its effect on a comparison of continuouswave (CW) light therapy versus pulsed wave (PW) treatments on SCI. Therats were transcutaneously irradiated within 15 minutes of SCI surgerywith an 808 nm (infrared) diode laser for 50 minutes daily andthereafter for 14 consecutive days. After an extended discussion, theauthors reported: “In conclusion, CW and pulsed laser light supportaxonal regeneration and functional recovery after SCI. Pulsed laserlight has the potential to support axonal regrowth to spinal cordsegments located farther from the lesion site. Therefore, the use ofpulsed light is a promising non-invasive therapy for SCI.”

While the majority of these studies utilized pulsed lasers, similarsystems were subsequently developed using digitally pulsedlight-emitting diodes (LEDs). These studies (e.g. Laser Med. Sci., 2009)showed that, all things being equal, LED phototherapy matches oroutperforms laser phototherapy. Moreover, LED therapy solutions arecheaper to implement and intrinsically offer greater safety than lasermethods and apparatus. Given these considerations, this applicationshall focus on LED based systems, but with the caveat that many of thedisclosed inventive methods are equally applicable for both LED orsemiconductor-laser based solutions.

Pulsed LED Phototherapy Systems

FIG. 1 illustrates elements of a phototherapy system capable ofcontinuous or pulsed light operation including an LED driver 1controlling and driving LEDs as a source of photons 3 emanating from LEDpad 2 on tissue 5 for the patient. Although a human brain is shown astissue 5, any organ, tissue or physiological system may be treated usingphototherapy. Before and after, or during treatment, doctor or clinician7 may adjust the treatment by controlling the settings of LED driver 1in accordance with monitor observations.

While there are many potential mechanisms, as shown in FIG. 2, it isgenerally agreed that the dominant photobiological process 22responsible for photobiomodulation during phototherapy treatment occurswithin a mitochondrion 21, an organelle present in every eukaryotic cell20 comprising both plants and animals including birds, mammals, horses,and humans. To the present understanding, photobiological process 22involves a photon 23 impinging, among others, a molecule cytochrome-coxidase (CCO) 24, which acts as a battery charger increasing thecellular energy content by transforming adenosine monophosphate (AMP)into a higher energy molecule adenosine diphosphate (ADP), andconverting ADP into an even higher energy molecule adenosinetriphosphate (ATP). In the process of increasing stored energy in theAMP to ADP to ATP, charging sequence 25, cytochrome-c oxidase 24 actssimilar to that of a battery charger with ATP 26 acting as a cellularbattery storing energy, a process which could be considered animal“photosynthesis”. Cytochrome-c oxidase 24 is also capable of convertingenergy from glucose resulting from digestion of food to fuel in the ATPcharging sequence 25, or through a combination of digestion andphotosynthesis.

To power cellular metabolism, ATP 26 is able to release energy 29through an ATP-to-ADP-to-AMP discharging process 28. Energy 29 is thenused to drive protein synthesis including the formation of catalysts,enzymes, DNA polymerase, and other biomolecules.

Another aspect of photobiological process 22 is that cytochrome-coxidase 24 is a scavenger for nitric oxide (NO) 27, an importantsignaling molecule in neuron communication and angiogenesis, the growthof new arteries and capillaries. Illumination of cytochrome-c oxidase 24in cells treated during phototherapy releases NO 27 in the vicinity ofinjured or infected tissue, increasing blood flow and oxygen delivery tothe treated tissue, accelerating healing, tissue repair, and immuneresponse.

To perform phototherapy and stimulate cytochrome-c oxidase 24 to absorbenergy from a photon 23, the intervening tissue between the light sourceand the tissue absorbing light cannot block or absorb the light. Theelectromagnetic radiation (EMR) molecular absorption spectrum of humantissue is illustrated in a graph 40 of absorption coefficient versus thewavelength of electromagnetic radiation λ (measured in nm) as shown inFIG. 3. FIG. 3 shows the relative absorption coefficient of oxygenatedhemoglobin (curve 44 a), deoxygenated hemoglobin (curve 44 b),cytochrome c (curves 41 a, 41 b), water (curve 42) and fats and lipids(curve 43) as a function of the wavelength of the light. As illustrated,deoxygenated hemoglobin (curve 44 b) and also oxygenated hemoglobin,i.e. blood, (curve 44 a) strongly absorb light in the red portion of thevisible spectrum, especially for wavelengths shorter than 650 nm. Atlonger wavelengths in the infrared portion of the spectrum, i.e. above950 nm, EMR is absorbed by water (H₂O) (curve 42). At wavelengthsbetween 650 nm to 950 nm, human tissue is essentially transparent asillustrated by transparent optical window 45.

Aside from absorption by fats and lipids (curve 43), EMR comprisingphotons 23 of wavelengths λ within in transparent optical window 45, isdirectly absorbed by cytochrome-c oxidase (curves 41 aa, 41 b).Specifically, cytochrome-c oxidase 24 absorbs the infrared portion ofthe spectrum represented by curve 41 b unimpeded by water or blood. Asecondary absorption tail for cytochrome-c oxidase (curve 41 a)illuminated by light in the red portion of the visible spectrum ispartially blocked by the absorption properties of deoxygenatedhemoglobin (curve 44 b), limiting any photobiological response for deeptissue but still activated in epithelial tissue and cells. FIG. 3 thusshows that phototherapy for skin and internal organs and tissue requiresdifferent treatments and light wavelengths, red for skin and infraredfor internal tissue and organs.

Present Photonic Delivery Systems

In order to achieve maximum energy coupling into tissue duringphototherapy, it is important to devise a consistent delivery system forilluminating tissue with photons consistently and uniformly. While earlyattempts used filtered lamps, lamps are extremely hot and uncomfortablefor patients, potentially can burn patient and doctors, and areextremely difficult in maintaining uniform illumination during atreatment of extended durations. Lamps also suffer short lifetimes, andif constructed using rarified gasses, can also be expensive to replaceregularly. Because of the filters, the lamps must be run very hot toachieve the required photon flux to achieve an efficient therapy inreasonable treatment durations. Unfiltered lamps, like the sun, actuallydeliver too broad of a spectrum and limit the efficacy of the photons bysimultaneously stimulating both beneficial and unwanted chemicalreactions, some involving harmful rays, especially in the ultravioletportion of the electromagnetic spectrum.

As an alternative, lasers have been and continue to be employed toperform phototherapy. Like lamps, lasers risk burning a patient, notthrough heat, by exposing tissue to intense concentrated optical power.To prevent that problem, special care must be taken that laser light islimited in its power output and that unduly high current producingdangerous light levels cannot accidentally occur. A second, morepractical problem arises from a laser's small “spot size”, theilluminated area. Because a laser illuminates a small focused area, itis difficult to treat large organs, muscles, or tissue and it is mucheasier for an overpower condition to arise.

Another problem with laser light results from its “coherence,” theproperty of light preventing it from spreading out, making it moredifficult to cover large areas during treatment. Studies reveal there isno inherent extra benefit from phototherapy using coherent light. Forone thing, bacterial, plant and animal life evolved on and naturallyabsorbs scattered, not coherent light because coherent light does notoccur naturally from any known light sources. Secondly, the first twolayers of epithelial tissue already destroy any optical coherence, sothe presence of coherence is really relegated to light delivery but notto its absorption.

Moreover, the optical spectrum of a laser is too narrow to fully exciteall the beneficial chemical and molecular transitions needed for toachieve high efficacy phototherapy. The limited spectrum of a laser,typically a range of ±3 nm around the laser's center wavelength value,makes it difficult to properly excite all the beneficial chemicalreactions needed in phototherapy. It is difficult to cover a range offrequencies with a narrow bandwidth optical source. For example,referring again to FIG. 3, clearly the chemical reactions involved inmaking the CCO absorption spectra (curve 41 b) is clearly different thanthe reactions giving rise to absorption tail (curve 41 a). Assuming theabsorption spectra of both regions are shown to be beneficial it isdifficult to cover this wide range with an optical source having awavelength spectrum only 6 nm wide.

So just as sunlight is an excessively broad spectrum, photobiologicallyexciting many competing chemical reactions with many EMR wavelengths,some even harmful, laser light is too narrow and does not stimulateenough chemical reactions to reach full efficacy in phototherapeutictreatment. This subject is discussed in greater detail in a relatedapplication entitled “Phototherapy System And Process Including DynamicLED Driver With Programmable Waveform”, by Williams (U.S. applicationSer. No. 14/073,371), now U.S. Pat. No. 9,877,361, issued Jan. 23, 2018,incorporated herein by reference.

To deliver phototherapy by exciting the entire range of wavelengths inthe transparent optical window 45, i.e. the full width fromapproximately 650 nm to 950 nm, even if four different wavelength lightsources are employed to span the range, each light source would requirea bandwidth almost 80 nm wide. This is more than an order of magnitudewider than the bandwidth of a laser light source. This range is simplytoo wide for lasers to cover in a practical manner. Today, LEDs arecommercially available for emitting a wide range of light spectra fromthe deep infrared through the ultraviolet portion of the electromagneticspectrum. With bandwidths of ±30 nm to ±40 nm, it is much easier tocover the desired spectrum with center frequencies located in the red,the long red, the short near infrared (NIR) and the mid NIR portions ofthe spectrum, e.g. 670 nm, 750 nm, 825 nm, and 900 nm.

FIG. 4 illustrates a preferred solution to light delivery problem is toemploy a flexible LED pad, one that curves to a patient's body as shownin pictograph 59. As shown, flexible LED pad 50 is intentionally bent tofit a body appendage, in this case leg comprising tissue 61, and pulledtaught by Velcro strap 57. To prevent slippage, flexible LED pad 50includes Velcro strips 58 glued to its surface. In use, Velcro strap 57wrapped around the pad attaches to the Velcro strips 58 holding flexibleLED pad 50 firmly in position conforming to a patient's leg, arm, neck,back, shoulder, knee, or any other appendage or body part comprisingtissue 61.

The resulting benefit, also shown in FIG. 4 illustrates that theresulting light penetration depth 63 into subdermal tissue 62 from LEDs52 embedded in flexible pad 50 is perfectly uniform along the lateralextent of the tissue 62. Unlike devices where the light source is astiff LED wand or inflexible LED panel held above the tissue beingtreated, in this example the flexible LED pad 50 is positioned adjacentto the patient's skin, i.e. epithelial 61, separated from the skin onlyby a disposable aseptic sanitation barrier 51, typically a clearhypoallergenic biocompatible plastic layer, which prevents theinadvertent spread of virulent agents through contact with LED pad 50.Close proximity between the LEDs 52 and the tissue 62 is essential tomaintain consistent illumination for durations of 20 minutes to over 1hour, an interval too long to hold a device in place manually. This isone reason handheld LED devices and gadgets, including brushes, combs,wands, and torchlights, have been shown to offer little or no medicalbenefit for phototherapy treatment.

A prior art phototherapy system for controlled light delivery availabletoday and shown in the pictograph of FIG. 5 comprises an electronicdriver 70 connected to one or more sets of flexible LED pads 71 a-71 ethrough cables 72 a and 72 b and connected to one other through shortelectrical connectors 73 a-73 d.

Specifically, one electrical output of electronic LED driver 70 isconnected to center flexible LED pad 71 a by electrical cable 72 a,which is in turn connected to associated side flexible LED pads 71 b and71 c through electrical connectors 73 a and 73 b, respectively. A secondset of LED pads connected to a second electrical output of electronicdriver 70 is connected to center flexible LED pad 71 c by electricalcable 72 b, which is in turn connected to associated side flexible LEDpads 91 d and 91 e through electrical connectors 73 c and 73 d,respectively, located on the edge of LED pad 71 c perpendicular to theedge where electrical cable 72 b attaches. The use of flexible LED padsand the ability of electronic LED driver 70 to independently drive twosets of LED pads with up to 900 mA of current, with each comprising aset of three pads, renders the phototherapy system a best-in-classproduct offering today.

Despite its technical superiority, the prior art phototherapy systemsuffers from numerous limitations and draw backs, including poorreliability for its LED pads, the inability to control LED current (andtherefore light uniformity) across the LED pads, limited control in theexcitation patterns driving the LEDs, limited safety and diagnosticfeatures, and the inability to communicate or receive updates via theinternet, wirelessly, or by cloud services. These various inadequaciesare addressed by a number of related patents.

Improving the reliability of the flexible LED pads is addressed indetail in a related application entitled “Improved Flexible LED LightPad for Phototherapy,” by Williams et. al. (U.S. application Ser. No.14/460,638) now U.S. Pat. No. 9,895,550, issued Feb. 20, 2018, which isincorporated herein by reference. FIG. 6A illustrates a view of theimproved flexible LED pad set, which virtually eliminates all discretewires and any wires soldered directly into PCBs within the LED pads(except for those associated with center cable 82) while enablingsignificantly greater flexibility in positioning and arranging theflexible LED pads upon a patient undergoing phototherapy.

As shown, the LED pad set includes three flexible LED pads comprisingcenter flexible LED pad 80 a with associated electrical cable 82, andtwo side flexible LED pads 80 b and 80 c. All three LED pads 80 a-80 cinclude two connector sockets 84 for connecting pad-to-pad cables 85 aand 85 b. Although connector socket 84 is not visible in thisperspective drawing as shown, its presence is easily identified by thehump 86 in the polymeric flexible LED pad 80 b, and similarly inflexible LED pads 80 a and 80 c. Pad-to-pad cables 85 a and 85 belectrically connect center LED pad 80 a to LED pads 80 b and 80 c,respectively.

Industry standard USB connectors maintain high performance andconsistent quality at competitive costs manufactured through a wellestablished high-volume supply chain, using sockets 84 that securelymount to a printed circuit board, and USB cables 85 a and 85 b, therebyintegrating electrical shielding and molded plugs and resisting breakagefrom repeated flexing and bending. Moreover, the USB connector cables 85a and 85 b are capable of reliably conducting up to 1 A of current andavoid excessive voltage drops or electromigration failures duringextended use. Aside from USB cables, other connector and cable setoptions include min-USB, IEEE-1394, and others. In the example shown inFIG. 6A, an 8-pin rectangular USB connector format was chosen for itsdurability, strength, and ubiquity.

In the embodiment shown in FIG. 6A, center flexible LED pad 80 a isrectangular and includes a strain relief 81 for connecting to cable 82and two USB sockets 84, all located on the same edge of center LED pad80 a, shown as the pad edge parallel to the x-axis. Similarly, each ofside LED pads 80 b and 80 c is also rectangular and includes two USBsockets also located on the same edge. This connection scheme ismarkedly different from the prior art device shown in FIG. 5, where theconnector sockets are proprietary and located on edges of the LED pads71 a-71 c and 71 c-71 e that face one another.

The benefit of this design change greatly improves a physician's orclinician's choices in positioning the LED pads on a patient beingtreated. Because the connector sockets do not face one another as theydo in prior art devices, connector cables 85 a and 85 b need not beshort in order to allow close placement of the LED pads. In fact, in theexample shown, LED pads 80 a, 80 b and 80 c may, if desired, actuallyabut one another without putting any stress on the cables 85 a and 85 bwhatsoever, even if long cables are employed. With the LED padstouching, the versatility of the disclosed flexible LED pad set offers adoctor the ability to utilize the highest number of LEDs in the smallesttreatment area.

Alternatively, the flexible LED pads may be placed far apart, forexample across the shoulder and down the arm, or grouped with two padspositioned closely and the third part positioned farther away. Withelectrical shielding in cables 85 a and 85 b, the pads may be positionedfar apart without suffering noise sensitivity plaguing the prior artsolutions shown previously.

The design shown in FIG. 6A also makes it easy for a clinician toposition the flexible LED pads 80 a-80 c, bend them to fit to thepatient's body, e.g. around the stomach and kidneys, and then secure thepads 80 a-80 c by Velcro belt 93 attaching to Velcro straps 92 attachedfirmly to the LED pads 80 a-80 c. The bending of the individual flexibleLED pads 80 a-80 c and the Velcro belt 93 binding them together isillustrated in FIG. 6B, where the belt 93 and the pads 80 a-80 c arebent to fit around a curved surface with curvature in the direction ofthe x-axis. In order to bend in the direction of the x-axis, no rigidPCB oriented parallel to the x-axis can be embedded within any of theLED pads 80 a-80 c.

In center LED pad 80 a, cable 82 and an RJ45 connector 83 are used toelectrically connect the LED pads 80 a-80 c to the LED controller inorder to preserve and maintain backward compatibility with existing LEDcontrollers operating in clinics and hospitals today. If an adapter forconverting RJ45 connector 83 to a USB connector is included, flexibleLED pad 80 a may be modified to eliminate cable 82 and strain relief 81,instead replacing the center connection with a third USB socket 84 andreplacing cable 82 with another USB cable similar to USB cable 85 a buttypically longer in length.

Methods of controlling LED current to improve light uniformity,providing enhanced safety and self-diagnostic capability whileaugmenting control of LED excitation patterns are described in theabove-referenced U.S. Pat. No. 9,877,361.

Control of LED Excitation Patterns

To precisely control the excitation pattern of the light pulses requiresmore sophisticated phototherapy system comprising advanced electroniccontrol. Such circuitry can be adapted from pre-existing driverelectronics, e.g. that used in HDTV LED backlight systems, re-purposedfor application to phototherapy.

As shown in FIG. 7, one such advanced electronic drive system adaptedfrom LED TV drive circuitry employs individual channel current controlto insure that the current in every LED string is matched regardless ofLED forward conduction voltages. As shown, current sinks 96 a, 96 b, . .. , 96 n are coupled to power N LED strings 97 a, 97 b, . . . , 97 n,respectively, acting as switched constant current devices havingprogrammable currents when they are conducting and the ability to turnon and off any individual channel or combination thereof dynamicallyunder control of digital signals 98 a, 98 b, . . . , 98 n, respectively.The number n can be any number of channels that are practical.

As shown, controlled current in current sink 96 a is set relative to areference current 99 at a magnitude Iref and maintained by a feedbackcircuit monitoring and adjusting the circuit biases accordingly in orderto maintain current I_(LEDa) in the string of m series-connected LEDs 97a. The number m can be any number of LEDs that are practical. Thecurrent control feedback is represented symbolically by a loop andassociated arrow feeding back into current sink 96 a. The digital enablesignals are then used to “chop” or pulse the LED current on and off at acontrolled duty factor and, as disclosed in the above-referenced U.S.Pat. No. 9,877,361, also at varying 27-29 frequencies. An LED controller103 is powered by low-dropout (LDO) linear regulator 102 and instructedby a microcontroller 104 through a SPI digital interface 105. A switchmode power supply 100 powers LED strings 97 a-97 n at a high voltage+V_(LED) which may be fixed or varied dynamically.

Despite employing analog current control, the resulting waveforms, andPWM control are essentially digital waveforms, i.e. a string ofsequential pulses as shown in FIG. 8A, controlling the average LEDbrightness and setting the excitation frequency by adjusting therepetition rate and LED on-times. As shown in the simplified timingdiagram of FIG. 8A, a string of clock pulses is used to generate asequential waveform of LED light, which may comprise differentwavelength LEDs of wavelengths λ_(a), λ_(b), and λ_(c), each illuminatedat different times and different durations.

As shown by the illustrative waveforms 110 and 111 in FIG. 8A, a pulsegenerator within LED controller 103 generates clock pulses at intervalsT_(θ) and a counter located within LED controller 103 associated withgenerating the waveform 111 counts 9 clock pulses and then turns on thespecific channel's current sink and λ_(a) LED string for a duration of 4pulses before turning it off again. As shown by waveform 112, a secondcounter, also within LED controller 103, turns on the λ_(b) channelimmediately after one clock pulse for a duration of 8 clock pulses, andthen turns the channel's LED string off for a duration of 4 clock pulses(while the λ_(a) LED string is on) and then turns the λ_(b) LED stringon again for another 3 clock pulses thereafter. As shown by waveform113, a third counter in LED controller 103 waits 22 pulses beforeturning on the λ_(c) LED string for a duration of 4 pulses then offagain.

In this sequenced manner, λ_(b) LED string conducts for a durationΔ_(t1) (8 clock pulses), then λ_(a) LED string conducts for a durationΔ_(t2) (4 clock pulses), then when it turns off λ_(b) LED stringconducts for a duration Δ_(t3) (3 clock pulses), waiting for a durationΔ_(t4) when no LED string is conducting, and followed by λ_(c) LEDstring conducting for a duration Δ_(t5) (4 clock pulses). The timingdiagrams 110-113 illustrate the flexibility of the new control system invarying the LED wavelength and the excitation pattern frequency.

The improved LED system allows precise control of the duration of eachlight pulse emitted by each of LED strings λ_(a), λ_(b) and λ_(c). Inpractice however, biological systems such as living cells cannot respondto single sub-second pulses of light, so instead one pattern comprisinga single wavelength and a single pattern frequency of pulses is repeatedfor long durations before switching to another LED wavelength andexcitation pattern frequency. A more realistic LED excitation pattern isshown in FIG. 8B, where the same clock signal (waveform 110) is used tosynthesize, i.e. generate, a fixed frequency excitation pattern 116 of asingle λ_(a) wavelength light with an synthesized pattern frequency off_(synth), wheref _(synth)=1/nT _(θ),where the time T_(θ) is the time interval at which successive clockpulses are generated, and “n” is the number of clock pulses in eachperiod of the synthesized waveform. As shown in waveform 116, until timet₁ the LED string is on 50% of the time so the duty factor D is 50% andthe brightness of the LED is equal to one-half of what it would be if itwere on all the time. After the time t₂, the duty factor is increased to75%, increasing average LED brightness but maintain the same synthesizedpattern frequency f_(synth).

Timing diagram 117 illustrates a similar synthesized waveform of asingle λ_(a) wavelength light at a fixed brightness and duty factorD=50% until time t₁. However, instead of varying the brightness at timet₂, the synthesized pattern frequency changes from f_(synth1)=1/nT_(θ)to a higher frequency f_(synth2)=1/mT_(θ), m being less than n. So attime t₂, the synthesized frequency increases from f_(synth1) tof_(synth2), even though the duty factor (50%) and LED brightness stayconstant. In summary, the improved LED drive system allows thecontrolled sequencing of arbitrary pulse strings of multiple and varyingwavelength LEDs with control over the brightness and the duration anddigital repetition rate, i.e. the excitation or pattern frequency.

To avoid any confusion, it should noted that the pattern frequencyf_(synth) is not the LED's light frequency. The light's frequency, i.e.the color of the emitted light, is equal to the speed of light dividedby the light's wavelength λ, or mathematically asυ_(EMR) =c/λ≈(3·10⁸ m/s)/(0.8·10⁻⁶ m)=3.8·10¹⁴ cycles/s=380 THzFor clarity's sake, the light's frequency as shown is referred to by theGreek letter nu or “υ” and not by the small letter f or f_(synth). Ascalculated, the light's electromagnetic frequency is equal to hundredsof a THz (i.e. tera-Hz) while the synthesized pattern frequency of thedigital pulses f_(synth) is general in the audio or “sonic” range (andat most in the ultrasound range) i.e. below 100 kHz, at least nineorders-of-magnitude lower. Unless noted by exception, throughout theremainder of this application we shall refer to the “color” of lightonly by its wavelength and not by its frequency. Conversely, the pulserate or excitation pattern frequency f_(synth) shall only be describedas a frequency and not by a wavelength.

Summary of Limitations in Prior-Art Phototherapy

Prior-art phototherapy apparatus remain limited by a number offundamental issues in their design and implementation including

-   -   use of lasers (instead of LEDs) limited by their intrinsically        narrow bandwidth of emitted light unable to simultaneously        stimulate the required range of chemical reactions necessary to        maximize photobiostimulation and optimize medical efficacy,    -   safety concerns in the use of lasers    -   LEDs mounted in a rigid housing unable to conform to treatment        areas    -   poor, improper, or ineffective modulation of phototherapy        excitation patterns        The last subject, ineffective modulation of phototherapy        excitation patterns represents a major challenge and opportunity        for improving photobiomodulation and treatment efficacy, one        which represents the focus of this disclosure.

BRIEF SUMMARY OF THE INVENTION

In accordance with this invention, the intensity of light used inphototherapy is varied gradually and repeatedly with regular periodicityrather than being administered as a series of square-wave pulses thatare either ON or OFF. In many embodiments the light is generated bystrings of light-emitting diodes (LEDs), but in other embodiments othertypes of light sources, such as semiconductor lasers, may be used. In apreferred embodiment, the light is sometimes varied in accordance with asingle frequency sinusoidal function, or a “chord” having two or moresine waves as components, but it will become apparent that thetechniques described herein can be employed to generate an infinitevariety of intensity patterns and functions.

In one group of embodiments, the intensity of light emitted by a stringof LEDs is varied by analogically controlling the gate voltage of acurrent-sink MOSFET connected in series with the LEDs. A gate drivercompares the current in the LED string against a sinusoidal referencecurrent, and the gate voltage of the current-sink MOSFET isautomatically adjusted by circuitry within the MOSFET driver until theLED and reference currents match and the LED current is at its desiredvalue. In this way, the LED current mimics the sinusoidal referencecurrent. The sinusoidal reference current can be generated in a varietyof ways; for example, with an LC or RC oscillator, a Wien bridgeoscillator or a twin T oscillator.

In an alternative version of these embodiments, the gate voltage of thecurrent-sink MOSFET is varied using a digital-to-analog (D/A) converter.The D/A converter is supplied with a series of digital values thatrepresent the values of a sine wave at predetermined instants of time,e. g. 24 values in a full 360° cycle. The digital values may representnot only a sine wave but also may be generated by or from a CD or DVD.

In a second group of embodiments, the LED current is controlleddigitally, preferably using pulse-width modulation (PWM). As in theprevious embodiment, a sine wave is broken down into a series of digitalvalues that represent its level at particular intervals of time. Theseintervals are referred to herein as having a duration T_(sync). A pulseis generated for each T_(sync) interval, its width representing thevalue of the sine wave in that interval. To do this, each T_(sync)interval is further broken down into a number of smaller intervals (eachhaving a duration referred to herein as T_(θ)), and the gate of thecurrent-sink MOSFET is controlled such that the LED current is allowedto flow during a number of these smaller T_(θ) intervals that representthe value of the sine wave. Thus, the current-sink MOSFET is turned ONfor part of each T_(sync) interval and turned OFF during the remainderof each T_(sync) interval. As a result, the level of the LED current isaveraged (smoothed out) into the form of a sine wave.

The gate of the current-sink MOSFET may be controlled by a precisiongate bias and control circuit that receives reference current from areference current source and an enable signal from a digitalsynthesizer. The digital synthesizer contains a counter that is set to anumber representative of the number of small T_(θ) intervals duringwhich the current-sink MOSFET is to be turned ON. The current-sinkMOSFET is turned ON, and the counter counts down to zero. When thecounter reaches zero, the current-sink MOSFET is turned OFF. Thecurrent-sink MOSFET remains OFF for a number of T_(θ) intervals equal tothe total number of T_(θ) intervals in a T_(sync) interval less thenumber of T_(θ) intervals during which the current-sink MOSFET wasturned on.

At the beginning of the next T_(sync) interval, a new numberrepresentative of the next value of the sine wave is loaded into thecounter in the precision gate bias and control circuit, and the processis repeated.

Controlling the LEDs in accordance with a sinusoidal function eliminatesthe harmonics that are produced when the LEDs are pulsed ON and OFFaccording to a square wave function, many of which may fall within the“audible” spectrum (generally less than 20,000 Hz) and may havedeleterious effects on a phototherapy treatment. Using the technique ofthis invention, the frequencies of the smaller intervals used inproducing the sinusoidal function (1/T_(sync) and 1/T_(θ)) can typicallybe set at above 20,000 Hz, where they generally have little effect onphototherapy treatments.

Chords containing multiple sinusoidal functions may be generated byadding the values of the component sine waves together. With the analogtechnique, the sine waves may be added together with an analog mixer, ora chord may be generated using a polyphonic analog audio source in lieuof an oscillator. With the digital technique, the numerical valuesrepresenting the component sine waves may be added together using anarithmetic logic unit (ALU). Another way of creating a chord is tocombine an analog synthesized waveform with a second digital pulsefrequency by “strobing” the analog waveform ON and OFF at a strobefrequency. The strobe frequency may be either higher or lower than thefrequency of the analog waveform. The strobe pulse may be generated byfeeding an analog sine wave to a divide by 2, 4 or 8 counter to producea second waveform 1, 2 or 3 octaves above the analog sine wave,respectively.

An advantage of using a D/A converter to generate an analog voltage orusing the digital technique is that treatment sequences (e.g., forparticular organs or tissues) may be stored digitally in a memory (e.g.,an EPROM) for convenient retrieval and use by a doctor or otherclinician.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a simplified pictorial representation of a phototherapytreatment.

FIG. 2 is a simplified pictorial representation of photobiomodulation ofcellular mitochondria.

FIG. 3 is a graph showing the absorption spectra of cytochrome-c (CCO),blood (Hb), water and lipids.

FIG. 4 is a photographic example and schematic representation of a LEDpad being used in a phototherapy treatment.

FIG. 5 is a view of a phototherapy system comprising a controller andsix flexible polymeric LED pads.

FIG. 6A is a schematic representation of a set of three flexiblepolymeric LED pads connected together and attached to a Velcro strap.

FIG. 6B is a schematic representation of the set of flexible polymericLED pads shown in FIG. 6A, bent slightly to conform to a patient's body.

FIG. 7 is an electrical schematic diagram of a current controlled LEDpulsed phototherapy system.

FIG. 8A is an exemplary timing diagram, showing the sequential pulsedexcitation of multiple wavelength LEDs with varying durations.

FIG. 8B is an exemplary timing diagram, showing the sequential pulsedexcitation of multiple wavelength LEDs with various combinations of dutyfactor and frequency.

FIG. 9A illustrates the time domain and Fourier frequency domainrepresentation of a digital (square wave) pulse.

FIG. 9B illustrates a discrete Fourier transform representation usingvarying numbers of summed sine waves.

FIG. 9C illustrates the measured current harmonic content of a digitallypulsed power supply.

FIG. 9D illustrates a measured Fourier spectrum of amplitude harmonics.

FIG. 9E illustrates a Fourier transform of a limited time sample of ameasured amplitude data revealing the frequency “spurs” resulting fromthe short duration sample.

FIG. 9F illustrates the magnitude of odd and even harmonics and thecumulative energy over the spectrum of a continuous Fourier transform ofa digital (square wave) pulse.

FIG. 10 illustrates a graph of the frequency response of an oscillatorysystem having two resonant frequencies.

FIG. 11 illustrates the summation of two synchronized digital pulses ofvarying frequency.

FIG. 12A illustrates a graph of spectral content of a 292 Hz digitalpulse contaminating the audio spectrum to that of idealized octaves ofD4 in the same range.

FIG. 12B illustrates a graph showing that the spectral content of a4,671 Hz digital pulse mostly contaminates the ultrasonic spectrum.

FIG. 13 illustrates various physical mechanisms of photobiomodulation

FIG. 14 illustrates two equivalent circuits of a single channel LEDdriver with current control.

FIG. 15 illustrates various example combinations of reference currentand enable signals and the resulting LED current waveforms.

FIG. 16A schematically illustrates the problem of current sharing amongmultiple loads from a single reference current.

FIG. 16B schematically illustrates the use of transconductanceamplifiers for distributing a reference current among multiple loads.

FIG. 16C schematically illustrates one implementation of a controlledcurrent sink comprising a high voltage MOSFET and MOSFET driver circuitwith resistor trimming.

FIG. 16D schematically illustrates one implementation of a controlledcurrent sink comprising a high voltage MOSFET and MOSFET driver circuitwith MOSFET trimming.

FIG. 17A schematically represents the use of a fixed-frequency voltagesource to generate an oscillating current reference.

FIG. 17B schematically represents the use of an adjustable voltagesource to generate an oscillating reference current.

FIG. 17C schematically represents a frequency and voltage adjustablevoltage source comprising a Wien-bridge used to generate an oscillatingreference current.

FIG. 17D schematically represents a programmable level shift circuitusing a resistor ladder.

FIG. 18A schematically represents an implementation of a single-channelcurrent-controlled LED driver using a D/A converter to generate areference current.

FIG. 18B schematically represents an implementation of a D/A converterusing a resistor ladder.

FIG. 19A illustrates a 292 Hz sine wave synthesized from a D/Aconverter.

FIG. 19B illustrates the harmonic spectra of 292 Hz sine wavesynthesized using a D/A converter generated reference current.

FIG. 19C illustrates an expanded view of digital steps present in a 292Hz sine wave synthesized from a D/A converter generated referencecurrent.

FIG. 19C illustrates an expanded view of digital steps present in a18.25 Hz sine wave synthesized from a D/A converter generated referencecurrent.

FIG. 19D illustrates a portion of a 18.25 Hz sine wave comprising asequence of voltage changes occurring at a clock frequency of a D/Aconverter.

FIG. 19E illustrates the harmonic spectra of a 18.25 Hz sine wavesynthesized using a D/A converter generated reference current.

FIG. 20 illustrates various combinations of sinusoidal referencecurrents and resulting LED current waveforms.

FIG. 21 illustrates the sum of two sinusoidal waveforms and theresulting waveform.

FIG. 22A schematically illustrates the use of an analog mixer togenerate a polyphonic oscillatory reference current for phototherapy LEDdrive.

FIG. 22B schematically represents the use of an analog audio source togenerate a polyphonic reference current for a phototherapy LED drive.

FIG. 22C schematically represents the use of a digital audio source togenerate a polyphonic reference current for a phototherapy LED drive.

FIG. 23A illustrates the synthesized polyphonic waveform generated froma sinusoidal reference current and a higher frequency digital pulse.

FIG. 23B illustrates the polyphonic harmonic spectra generated from a292 Hz sinusoidal reference current and a 4,672 Hz digital pulse.

FIG. 23C illustrates the polyphonic harmonic spectra generated from a292 Hz sinusoidal reference current and a 9,344 Hz digital pulse.

FIG. 23D illustrates the polyphonic harmonic spectra generated from a292 Hz sinusoidal reference current and an ultrasonic digital pulse.

FIG. 23E illustrates the polyphonic harmonic spectra generated from a292 Hz sinusoidal reference current and a 18,688 Hz digital pulse.

FIG. 24 illustrates the synthesized polyphonic waveform generated from asinusoidal reference current and a lower frequency digital pulse.

FIG. 25A illustrates the polyphonic harmonic spectra generated from a9,344 Hz sinusoidal reference current and a 4,672 Hz digital pulse.

FIG. 25B illustrates the polyphonic harmonic spectra generated from a584 Hz sinusoidal reference current and a 292 Hz digital pulse.

FIG. 26 schematically illustrates implementation of a polyphonic LEDcurrent drive for phototherapy from a single oscillator.

FIG. 27A schematically illustrates multiple digital synthesizerscontrolling multiple corresponding LED drivers.

FIG. 27B schematically illustrates a centralized digital synthesizerseparately controlling multiple LED drivers.

FIG. 27C schematically illustrates a single digital synthesizercontrolling multiple LED drivers with a common signal.

FIG. 28A illustrates a circuit diagram of a digital synthesizer.

FIG. 28B is a timing diagram of digital synthesizer operation.

FIG. 28C illustrates synthesized pulses of a fixed frequency and varyingduty factor.

FIG. 29A illustrates an LED drive waveform comprising a fixed frequencyPWM synthesized sinusoid.

FIG. 29B illustrates examples of digitally synthesized sinusoids.

FIG. 29C illustrates a comparison of the output waveforms of a D/Aconverter versus PWM control over a single time interval.

FIG. 29D graphically illustrates interrelationship between PWM bitresolution, the number of time intervals, and the maximum frequencybeing synthesized to the required counter clock frequency.

FIG. 30 schematically illustrates a clock generator circuit.

FIG. 31 graphically illustrates the dependence of overall digitalsynthesis resolution and PWM bit resolution on the maximum frequencybeing synthesized.

FIG. 32A illustrates the frequency spectrum of a digitally synthesized4,672 Hz sinusoid.

FIG. 32B illustrates the frequency spectrum of a digitally synthesized292 Hz sinusoid.

FIG. 32C graphically illustrates the dependence of the Sync and PWMcounter frequencies on the synthesized frequency.

FIG. 33 illustrates a flow chart of sinusoidal waveform generation usingthe disclosed digital synthesis methods.

FIG. 34A graphically illustrates digital synthesis of a 292 Hz (D4) sinewave using 15° intervals.

FIG. 34B graphically illustrates digital synthesis of a 292 Hz (D4) sinewave using 20° intervals.

FIG. 34C graphically illustrates the PWM intervals used in the digitalsynthesis of a 292 Hz (D4) sine wave using 20° intervals.

FIG. 34D graphically illustrates the digital synthesis of a 1,168 Hz(D6) sine wave using 20° intervals.

FIG. 34E graphically illustrates the digital synthesis of a 4,672 Hz(D6) sine wave using 20° intervals.

FIG. 35A graphically illustrates the digital synthesis of a 1,168 Hz(D6) sine wave with a 50% amplitude.

FIG. 35B graphically illustrates the digital synthesis of 1,168 Hz (D6)sine wave with a 50% amplitude offset by +25%.

FIG. 35C graphically illustrates the digital synthesis of a 1,168 Hz(D6) sine wave with a 20% amplitude offset by +60%.

FIG. 35D illustrates the frequency spectrum of a digitally synthesized1,168 Hz (D6) sinusoid with a 20% amplitude offset by +60%.

FIG. 36 graphically illustrates the digital synthesis of 4-cycles of a4,472 Hz (D8) sine wave using 20° intervals.

FIG. 37A graphically illustrates the digital synthesis of a 1,168 Hz(D6) sine wave using 4× oversampling.

FIG. 37B illustrates the pattern file for the digital synthesis of a1,168 Hz (D6) sine wave using 4× oversampling.

FIG. 38 graphically illustrates the digital synthesis of a chord of4,472 Hz (D8) and 1,1672 Hz (D6) sinusoids of equal amplitude.

FIG. 39 illustrates the frequency spectrum of digitally synthesizedchord of 4,472 Hz (D8) and 1,1672 Hz (D6) sinusoids of equal amplitude.

FIG. 40 graphically illustrates the digital synthesis of a chord of4,472 Hz (D8) and 1,1672 Hz (D6) sinusoids of differing amplitudes.

FIG. 41 illustrates an algorithm for generating a synthesis patternfile.

FIG. 42A illustrates an algorithm for generating chords of two or moresinusoids in real time or in advance for storage in a pattern library.

FIG. 42B illustrates an alternative way of creating chords utilizing thealgorithm described in FIG. 41 to generate individual sinusoidal patternfiles with normalized mathematical functions.

FIG. 43 illustrates sinusoids of frequencies that are integral multiplesof one another.

FIG. 44 illustrates sinusoids of frequencies that are fractionalmultiples of one another.

FIG. 45 illustrates the use of mirror phase symmetry to generate a chordconsisting of sinusoids whose frequencies have a ratio of 11.5.

FIG. 46 illustrates the use of an interpolated gap fill to generate achord consisting of sinusoids having frequencies that are in anirregular ratio (1.873) to one another.

FIG. 47 illustrates generating a sinusoid using PWM while varying thereference current βI_(ref).

FIG. 48 illustrates how a prior art digital pulse circuit used to driveLED strings may be repurposed for the synthesis of sinusoidal waveforms.

FIG. 49 illustrates various physiological structures and conditions thatmay be amenable to treatment with phototherapy, as a function of theamplitude, frequency and DC component of the sinusoidal current used toilluminate the LEDs.

DESCRIPTION OF THE INVENTION

Harmonic Spectra of Synthesized Patterns

As described previously, the pulsing of light at prescribed frequenciesin prior art phototherapy is based on empirical evidence and doctors'observations that pulsed laser light works better than continuous lightin reducing pain and healing tissue. As stated previously, while thisgeneral conclusion appears credible, no consensus exists on what digitalpulses produce the best results and highest treatment efficacies. Todate, studies of laser phototherapy did not consider arbitrary waveforms(such as sine waves, ramp waves, sawtooth waveforms, etc.) but wererestricted to direct comparisons between continuous wave laser operation(CW) to pulsed wave (PW) laser operation, i.e. square waves, likelybecause most lasers are designed only to operate by being pulsed on oroff digitally. The pulse rates used were chosen to operate at a ratenear the time-constants of specific, empirically-observedphotobiological processes, i.e. in the audio range below 20 kHz.

In these studies, experimenters report the digital pulse rate anderroneously assume that this square-wave pulse frequency used tomodulate the light is the only frequency present in the test. Fromcommunication theory, physics, electromagnetics, and the mathematics ofFourier, however, it is well known that digital pulses do not exhibitonly the digital pulse frequency, but in fact exhibit an entire spectrumof frequencies. So, while it may seem reasonable to assume digitalpulses operating at a fixed clock rate both emit and conduct only asingle frequency—the fundamental switching frequency, this self-evidenttruth is, in fact, incorrect.

In fact, the harmonic content in switched digital systems can besignificant both in energy and in the spectrum the harmonicscontaminate—some harmonics occurring at frequencies that are orders ofmagnitude higher than the fundamental frequency. In electromagnetics,these harmonics are often responsible for unwanted conducted andradiated noise, potentially adversely affecting circuit operatingreliability. At higher frequencies, these harmonics are known togenerate electromagnetic interference, or EMI, radiated into thesurroundings.

Mathematical analysis reveals that the speed of the digital on-and-offtransitions (along with any possible ringing or overshoot) determine thegenerated harmonic spectra of a waveform. In power electronic systemssuch as the LED or laser drivers used in phototherapy systems, theproblem is compounded by high currents, large voltages and high powerdelivered in such applications because more energy is being controlled.In fact. unless the precise rise time and fall time of a string ofdigital pulses is accurately recorded, the frequency spectrum resultingfrom the string of pulses is unknown.

The origin and magnitude of these unexpected frequencies can best beunderstood mathematically. Analysis of any physical system or anelectrical circuit may be performed in the “time domain”, i.e. wheretime is the key variable by which everything is measured and referenced,or alternatively in the “frequency domain”, where every time-dependentwaveform or function is considered as a sum of sinusoidal oscillatingfrequencies. In engineering, both time and frequency domains are usedinterchangeably, essentially because some problems are more easilysolved in the time domain and others are better analyzed as frequencies.

One means to perform this translation between time and frequency isbased on the 18^(th) century contributions of the French mathematicianand physicist Jean Fourier which revealed that generalized functions maybe represented by sums of simpler trigonometric functions, generallysine and cosine waveforms (a cosine may be considered as a sine waveshifted by 90° in phase). The methodology is bidirectional—Fourieranalysis comprises decomposing or “transforming” a function into itssimpler elements, or conversely, synthesizing a function from thesesimpler elements. In engineering vernacular, the term Fourier analysisis used to mean the study and application of both operations.

A continuous Fourier transform refers to the transform of a continuousreal argument into a continuous frequency distribution or vice versa.Theoretically, the continuous Fourier transform's ability to convert atime varying waveform into the precise frequency domain equivalent,requires summing an infinite number of sine waves of varying frequencyand sampling the time dependent waveform for an infinite period of time.An example of this transform is shown in FIG. 9A, where graph g(t)illustrates a repeating time dependent waveform 118. The equivalentfrequency domain spectrum is shown by graph G(f) illustrating that asimple square wave results in an continuous spectrum 119 of frequenciesof varying magnitudes centered around the fundamental frequency f=0.

Of course, taking data samples for infinite time and summing an infinitenumber of sine waves are both idealized impossibilities. In mathematicsand control theory, however, the word “infinite” can be safelytranslated into a “very large number”, or even more practically inengineering to mean a “large number compared to what is being analyzed”.Such an approximation of a series sum of a limited number of “discrete”sinusoids is referred to as a discrete Fourier transform or “Fourierseries”. In practice, measuring a regularly repeating time domainwaveform for 2 to 5 periods can be very accurately emulated with the sumof less than 50 sinusoids of varying frequencies. Moreover, in caseswhere the original time-domain waveform is simple, regular and repeatingfor extended duration, reasonable approximations can occur by summingonly a few sinusoids.

This principle is illustrated in FIG. 9B in a graph of signal magnitude,in this case LED current, versus time in four different casesapproximating square wave 117 using the discrete Fourier transformmethod. In the four cases shown the number of sine waves K used in thetransform vary from K=1 to K=49. Clearly, in the case of K=1, the singlesinusoid only vaguely resembles square wave 117. When the number of sinewaves of varying frequency used in the transform is increased to K=5,the resulting reconstructed waveform 121 and its match to square wave117 improves dramatically. At K=11, the match of waveform 121 veryclosely tracks the original 117, while at K=49 the transformreconstruction and the original waveform are nearly indistinguishable.

Through Fourier analysis, then, physicists can observe what frequenciesare present in any time varying system or circuit by looking at theconstituent components and the amount of energy present in eachcomponent. This principle is exemplified in the graph of FIG. 9C showingthe measured spectral components of current in a power circuitcomprising a 150 Hz square wave. The Fourier transform was performed bythe measuring device employing a real time analytical algorithm called aFFT or fast Fourier transform to immediately estimate the measuredspectra from a minimal data sample. As shown by spike 125, thefundamental pulse frequency is at 150 Hz and has an amplitude of 1.2 A.The fundamental frequency is accompanied by a series of harmonics at 450Hz, 750 Hz, 1050 Hz, and 1350 Hz, corresponding to the 3^(rd), 5^(th),7^(th) and 9^(th) harmonics of the fundamental frequency. The 9^(th)harmonic 127 has a frequency well into the kHz range despite the lowfundamental pulse rate. Also, it should be noted the 3^(rd) harmonic 126is responsible for 0.3 A of the current in the waveform, a substantialportion of the current flowing in the system. As shown, the circuit alsoincluded a 2.5 A DC component of current 128, i.e. at a frequency of 0Hz. A steady DC component does not contribute to the spectraldistribution and can be ignored in a Fourier analysis.

FIG. 9D illustrates another example of a FFT, this time with the signalamplitude measured in decibels (dB). As shown, the 1 kHz fundamental isaccompanied by a sizeable 3^(rd) harmonic 131 at 3 kHz and includesspectral contributions 132 above—30 dB beyond 20 kHz. In contrast, FIG.9E illustrates a less idealistic looking FFT output of a 250 Hz squarewave with a fundamental frequency 135 of 250 Hz, a 3^(rd) harmonic 136of 750 Hz, and a 15^(th) harmonic 137 of 3750 Hz. The lobes 138 aroundeach significant frequency and the inaccuracy of the frequency can becaused to be two phenomena, either a small and inadequate time basedsample measurement possibly with jitter in the signal itself, or thepresence of high frequency fast transients that do not appear in normaloscilloscope waveforms but distort the waveform. In this case, as inevery prior example shown, the FFTs of a square wave, i.e. a repeatingdigital pulse, exhibit purely odd harmonics of the fundamental.

The behavior of a square wave or a string of digital pulses issummarized in the discrete Fourier transform calculation of a squarewave shown in FIG. 9F, where the fundamental frequency 140 isaccompanied only by odd harmonics 141, 142, 143 . . . 144 correspondingto the 3^(rd), 5^(th), 7^(th), . . . , 19^(th) harmonics. All evenharmonics 145 of the fundamental frequency f₁ carry no energy, meaningtheir Fourier coefficient is zero, i.e. they do not exist. If the y-axisalso represents the cumulative current or energy of the fundamental andeach harmonic component, then assuming the total current is present inthe first 20 harmonics and all other harmonics are filtered out, thefundamental alone represents only 47% of the total current as shown bycurve 146. This means that less than half the current is oscillating atthe desired frequency. Including the 3^(rd) harmonic 141, the totalcurrent is 63%, while adding the 5^(th) and the 7^(th) increases thecontent to 72% and 79% respectively.

While even harmonics, e.g. 2^(nd), 4^(th), 6^(th), . . . , (2n)^(th),tend to reinforce their fundamental frequency, it is well known that oddharmonics tend to interfere, i.e. fight, with one another. In the audiospectrum, for example, vacuum tube amplifiers produce even harmonicdistortion, a sound that sounds good to the human ear. Bipolartransistors, on the other hand produce odd harmonics that interfere withone another, in the audio spectrum producing a scratchy uncomfortablesound, wasting energy. Whether these frequencies are exciting an audiomembrane, e.g. a speaker transducer, a microphone transducer, or thehuman ear drum, or whether they are exciting a molecule or a group ofmolecules, the result is the same—orderly oscillations of even harmonicsexhibit constructive interference enforcing the oscillations, whilecompeting random oscillations of odd harmonics result in random and eventime-varying waveforms manifesting destructive wave interference,producing erratic inefficient energy coupling in a system, and sometimeseven giving rise to unstable conditions in the system.

Such is the case in any physical system that can absorb and temporarilystore energy then release the energy kinetically. To understand theinteraction of such a physical system excited with a spectrum offrequencies, however, the concept of oscillatory behavior and resonancemust be considered. Thereafter, the behavior of chemical and biologicalsystems, which follow the same laws of physics, can more thoroughly beconsidered.

Principles of Oscillations and Resonance

In any physical system capable of manifesting both kinetic energy, i.e.the energy of motion, and potential energy, i.e. stored energy, thereexists the capability of oscillatory behavior and “resonance”.Oscillations occur when energy repeatedly transfers from one form ofpotential energy into another. In mechanical examples the compressionand expansion of a spring represent an oscillatory system where thespring's tension represents stored energy and where a swinging doorrepresent the kinetic energy of motion and its associated frictionleading to energy loss. A similar example is a pendulum or a childswinging in a swing, each time stopping at the top of each arc (wherekinetic energy is zero and potential energy is maximum) and then fallingback to earth as the swing reaches the bottom of its arc (where thepotential energy is at is minimum value and the velocity and kineticenergy of the swing is at its maximum value). In such an example thepotential energy is stored in the force due to gravity. Similarphenomena occur in buildings and bridges, sensitive to both wind andseismic vibrations. Each time the object oscillates friction removessome of the energy and the system loses its total energy. Unless thatenergy is replenished the system will eventually lose all of its energyand cease oscillating.

The mechanism of oscillatory behavior is also manifest in electricalcircuits with magnetic and capacitive elements, where the energy may bestored in a magnetic field, or an electric field or some combinationthereof. The current and voltage in inductive and capacitive elementsare intrinsically out of phase and once energized, spontaneouslyoscillate, with stored energy being redistributed from the inductor tothe capacitor, or vise versa. During the oscillations, whenever currentis flowing between the energy storage elements, some of the system'senergy is lost as heat as a result of electrical resistance.

At sufficiently high oscillating frequencies, however, the electricfield and magnetic field can no longer be contained within the circuitelements. The resulting electro-magnetic field propagates through spaceas an electromagnetic “traveling” wave, also known as electromagneticradiation or EMR. Depending on the oscillatory frequency, EMR maycomprise radio waves, microwaves, infrared radiation, light, ultravioletlight, X-rays, or gamma-rays. In the vacuum of space EMR can travelindefinitely. By contrast, for any EMR traveling through matter, thewave is gradually attenuated and energy is lost as it travels, in amanner similar to the energy loss due to friction in mechanical systemsor to losses due to resistance in electrical circuits.

In any system capable of exhibiting oscillatory behavior, the timing ofwhen energy is put into the system determines its response. In the swingexample, if an adult pushes the swing before it has returned fully tothe apex of its height, the pushing force will act against the swing'sswinging motion and reduce its energy lowering the maximum height towhich the swing reaches on its next cycle. The action of pushing tooearly impedes or interferes with the swing's motion and can be referredto as destructive interference. Conversely, if the adult waits tillafter the swing reaches its peak height where the swing reversesdirection, pushing at that time will put energy into the swing andreinforce the oscillation making the swing reach a higher height on itsnext oscillatory cycle. The action of pushing at just the right time,thereby reinforcing the swing's motion, can be referred to asconstructive interference. If the pushing is done cyclically at just theright moment the swing will go higher with each cycle and the benefitfrom pushing at the right time maximizes the energy transfer into theswing's oscillations. The swing is said to be oscillating near its“resonant” frequency.

The same thing is true in an electrical system. In a system RLCoscillatory circuit or RLC “tank”, energy “sloshes” back and forthbetween the inductor L and the capacitor C (hence the metaphor of watersloshing to and fro in a “tank”). If an oscillating source of energysuch as an AC voltage source driving the network oscillates with afrequency approaching the value 1/SQRT(LC), the oscillations will reachtheir maximum magnitude and the energy coupling from the AC voltagesource into the tank circuit will be greatest. The presence ofresistance R causes a loss in energy in the tank circuit. Any excitationfrequency below or above the resonant frequency will couple energy intothe circuit less efficiently than at the resonant frequency.

To better envision this behavior, the frequency of the oscillatingvoltage source exciting the oscillating tank circuit is swept startingfrom a low frequency below resonance up and increased constantly to ahigher value. At very low frequencies (near DC) the tank circuit may notreact at all. As the frequency ramps, energy couples into the system andcurrent begins to oscillate between the inductor and capacitor. As thedriving frequency continues to increase, the response of the tank to theexcitation and the corresponding magnitude of the oscillations willgrow, steadily at first and then rapidly as the resonant frequency isapproached. When the driving voltage source reaches the circuit'sresonant frequency the oscillations will hit their peak value and mostefficient energy transfer. Continuing to ramp the frequency beyondresonance will reduce the magnitude of the oscillations.

While the example cited describes a system with a single resonantfrequency, oftentimes a system contains more than two energy storageelements, conditions or mechanisms and may therefore exhibit two or morenatural resonant frequencies. An example of a system with two resonantfrequencies is shown in the graph of FIG. 10 as a plot of the magnitudeof an oscillation G(f) on y-axis and with frequency f on the x-axis. Asshown, response curve 151 includes a lower-frequency resonant peak 152at a frequency f1 and a second higher-frequency resonant peak 153 at afrequency f2. As shown, resonant peak 152 is greater in magnitude andbroader in frequency than resonant peak 153, which exhibits a lowermagnitude and a sharper sensitivity to frequency. The magnitude of thesystem's response between the two resonant peaks never reaches zero,meaning the entire system of energy storage elements are interacting atthose excitation frequencies.

So using the prior analytical method, sweeping a single AC voltagesource from low to high frequency, will trace out curve 151 startingwith an increase in G(f) until resonant peak 152 at frequency f1 isreached then declining and flattening at a lower magnitude until theresponse again grows as the driving frequency approaches f2 and resonantpeak 153, beyond which the response declines. In many instances,physical systems include resonant peaks that are never observed becausethey are never excited under normal conditions. A classic example ofthis behavior is a building that sways harmlessly at a fixed frequencyin the wind, but in an earthquake resonates severely at a lowerfrequency leading to building collapse. So in any oscillatory system, ifthe resonant frequencies are not known it is difficult to analyze asystem's response to excitation, especially unintended excitation.

Even worse, if the energy source providing the excitation itselfincludes a broad and unknown spectrum of frequencies, it is difficult topredict, understand, or even interpret the system's response. Such isthe problem with digital pulse excitation of an oscillatory system withmultiple resonant frequencies. Since each digital pulse generates afundamental frequency and a spectrum of harmonics, the variousfrequencies may stimulate unknown, unwanted, or even potentially harmfulharmonics.

In other cases, it may be desirable to intentionally stimulate severalspecific resonant frequencies but not others. In such cases, digitalpulses are also undesirable since the harmonics cover a range offrequencies and may stimulate unwanted resonances. Ideally then, it ispreferable in such circumstances to generate oscillations at the twotarget frequencies, e.g. at f1 and f2. Unfortunately, even ignoring theproblem of harmonics, another limitation of digital pulse control togenerate pulses at or near a desired frequency, is that the fundamentalexcitation frequency is intrinsically monophonic, i.e. comprising asingle frequency, pitch, or note.

For example, as shown in FIG. 11, if a system continuously generatesdigital pulses 191 at 60 Hz, and then we add a second series of digitalpulses 155 at 120 Hz on top of and synchronized to the original 60 Hzpulses 156, the resulting waveform 193 is identical to the digitalpulses 157 at 120 Hz with no 60 Hz component. This means for evenmultiples of synchronized digital pulses, only the highest frequencymultiple is manifest. In essence, when using digital pulses modulated ator near a desired frequency, it is only possible to excite a circuit oran energy conversion device (such as a laser or LED) with one singlefundamental frequency, so it is not possible to produce chords ormultiple frequencies simultaneously with the digital technology andmethods used in today's phototherapy apparatus.

Limitations of Pulsed Phototherapy

In conclusion, Fourier analysis reveals that using digital pulses tocontrol the brightness and pattern frequency of an electrical load, suchas an LED or a laser used in a phototherapy system, results in aspectrum of frequencies well beyond that of the fundamental frequencyused to pulse the energy conversion device. The resulting harmonicspectrum, comprising odd harmonics, wastes energy and potentiallycompromises a phototherapy device's ability to acutely control anddeliver a specific desired frequency of operation for an electroniccircuit or in an energy conversion device (such as a laser or LED).

Applying principles of oscillation and resonance to phototherapy,digital modulation of LED or laser light results in a broad spectrum offrequencies potentially exciting various chemical and photobiologicalprocesses in an uncontrolled manner. Since the frequencies needed toactivate particular chemical reactions in the healing process are notaccurately known, stimulating tissue with an uncontrolled spectrum ofharmonics renders identification and isolation of key beneficialfrequencies and the systematic improvement of treatment efficacyimpossible.

Along with ambiguity stemming from inadequately reported testconditions, harmonic spectral contamination resulting from square-wavepulsing of a light source during phototherapy experiments represents anuncontrolled variable responsible, at least in part, for the conflictingresults and inconsistent efficacies observed reported in publishedstudies attempting to optimize pulsed wave phototherapy. Assuming thatmost photobiological processes occur in the audio spectrum, i.e. below20 kHz, then analysis shows the impact of spectral contamination frompulsed operation should be worse at lower digital pulse frequenciesbecause unwanted harmonic spectrum generated more significantly overlapsand influences the frequencies sensitive to photobiological stimulation.

For example, the harmonic spectrum of a 292 Hz square wave pulsecontaminates most of the audio spectrum, while significant harmonicsgenerated from a 5 kHz square wave pulse occur in the ultrasonic range,i.e. >20 kHz, and beyond a cell's ability to react to such rapidfrequencies.

To elaborate on this point, FIG. 12A graphically contrasts the harmoniccontent of a 292 Hz digital pulse to that of a pure tone of 292 Hz, i.e.the fourth octave of D (or D4) and even multiples of this frequency, asrecommended by Nogier's studies on healing. Using pure tones, a 292 Hzfundamental frequency 161 would exhibit constructive interference andimproved energy transfer when blended with other harmonic multiples of Din the audio spectrum 163, for example D5, D6, D7, and D8 atcorresponding frequencies of 584 Hz, 1,168 Hz, 2,336 Hz, and 4,672 Hz.Instead, a 292 Hz repeating digital pulse 162 results in odd harmonics164 comprising 3^(rd), 5^(th), 7^(th), 9^(th), 11^(th), 13^(th),15^(th), . . . , harmonic frequencies at 876 Hz, 1,460 Hz, 2,044 Hz,2,628 Hz, 3,212 Hz, 3,796 Hz, 4,380 Hz and so on, none of which evenremotely match the even harmonic frequencies recommended byphysiological studies. Instead, the resulting spectrum content of oddharmonics 164 generated by 292 Hz digital pulse 162 contaminates much ofthe audio spectrum where adverse or non-beneficial interaction with manybiochemical processes may occur and interfere with desiredphotobiomodulation.

While digital pulses produce unwanted harmonics, not all pulsefrequencies should have an equally significant impact on biologicalprocesses and photobiomodulation. FIG. 12B contrasts a 4,672 Hz digitalpulse 172 and its generated odd harmonics 174 to a pure tone of D in theeighth octave 171 (i.e. D8), which also has a frequency of 4,672 Hz, andeven harmonics 173 of the pure tone D8. Specifically, a pure tone of Din the eighth octave 171 includes even multiples of this frequency, D9and D10 at 9,344 Hz and 18,688 Hz, respectively, in the audio rangewhere most photobiomodulation occurs. In contrast, at 37,376 Hz, thenote D11 is in the ultrasonic spectrum, a range of notes above thefrequency illustrated by line 175 that is too high to be heard and formost cells or tissue to react to chemically. The key point of this graphis that, despite the fact that a 4,672 Hz digital pulse 172 results in awhole spectrum of odd harmonics 174, only a single harmonic, the 3^(rd)harmonic at 14,016 Hz, falls within in the audio spectrum and below thefrequency specified by line 175. All the other harmonics are too high infrequency for most tissues to respond or react to significantly.

In conclusion, the spectral contamination resulting from digital pulsesis more significant at lower frequencies, because above 5 kHz pulserates, most of the unwanted odd harmonics that occur are ultrasonic,above the audio frequency range and at frequencies too high to adverselyimpact beneficial photobiomodulation.

Also, aside from producing undesirable harmonics, controlling a laser oran array of LEDs with a digital excitation pattern of pulses in adesired frequency range is incapable of producing chords or multiplefrequencies simultaneously, thereby limiting a phototherapy device'spotential for controlling or optimizing energy coupling into cells,tissue, or organs.

What is needed is a means to control the excitation pattern operation ofa laser or LED array to synthesize a specific desired frequency or groupof frequencies (chords) without spectral contamination from unwanted anduncontrolled harmonics, especially those contaminating the audiospectrum, i.e. below 20 kHz.

Improving Photobiomodulation Through Harmonic Control

In order to provide complete control of photobiomodulation duringphototherapy treatments (low-level light therapy or LLLT), the disclosedsystem described herein is capable of systematically driving arrays ofvarious wavelength LEDs or lasers with user-selectable arbitrarywaveforms (and sequences of waveforms) comprising continuous andtime-varying modulation patterns, frequencies and duty factors, free ofunwanted harmonics or spectral contamination. Time varying waveformscomprise digital pulses, sinusoids, pulsed sinusoids, continuousoperation, and user-defined waveforms and mathematical functions.

The goal of this enhanced control is to improve treatment efficacy byadjusting device operation to synchronize to natural frequencies ofparticular biological processes specific to cells, tissue, organs, andphysiological systems. By timing the energy delivery and controlling itsfrequencies and harmonics, tissue specificity can be enhanced. In orderto ascertain these operating parameters, the biochemical and cytologicalorigin of the frequency dependence of photobiomodulation must first beconsidered, starting with present-day knowledge and available technicalliterature.

Origin of Photobiomodulation Frequency Dependence

The frequency dependence of photobiomodulation and its influence onphototherapy efficacy is correlated to physical mechanisms within livingcells, tissue, organs, and physiological systems.

According to the previously cited paper, “Effect of Pulsing in Low-LevelLight Therapy” published in Lasers Surg. Med. August 2010, volume 42(6),pp. 450-466, “if there is a biological explanation of the improvedeffects of pulsed light it is either due to some fundamental frequencythat exists in biological systems in the range of tens to hundreds ofHz, or alternatively due to some biological process that has a timescale of a few milliseconds.”

This paper cites various natural frequencies occurring within livingorganisms, including electroencephalography studies identifying fourdistinct classes of brain waves, namely alpha waves at 8-13 Hz, betawaves at 14-40 Hz, delta waves at 1-3 Hz, and theta waves at 4-7 Hz.These various waves are present during different conditions or sleep,rest, meditation, visual, and cognitive mental activity and are affectedby illness, concussion and traumatic brain injury, and age. The authorssurmise “The possibility of resonance occurring between the frequency ofthe light pulses and the frequency of the brain waves may explain someof the results with transcranial LLLT using pulsed light.”

Similar observations have been made by other authors in regards toelectrocardiogram signals and regulation of heart function. Restingheart rates typically occur in the 60 to 100 beats per minute, orroughly 1 Hz to 2 Hz, depending on a person's age and health.Peristaltic contractions in the intestines can occur in sub 1 Hzfrequencies. These systems and their optimum response conditions do notrepresent simple chemical or electrical reaction rates because they areoperating as a clocked system with their own time regulation, generallyelectrochemical in nature. For example, through an electrochemicalprocess, potassium is intimately involved in setting the heart's naturalpulse rate in humans.

Another entirely different class of mechanisms present within cells andpotentially responsible for the photobiomodulation frequency dependenceappears related to chemical or ionic reaction rates and ionic transport.The Hashmi et al. paper continues, “the time scale for opening andclosing of ion channels is of the order of a few milliseconds,” withreferenced citations having time constants for ion channels ranging from0.1 to 100 milliseconds, i.e. 100 Hz to 10 Hz, including potassium andcalcium ion channels in mitochondria. Other papers suggest sarcolemma,the lipid bilayer plasma membrane providing scaffolding for musclecells, may also be responsible for photobiomodulation frequencydependence, since such membranes often serve as ion pumps.

On a cellular level, another mechanism responsible forphotobiomodulation frequency dependence is the photodissociation ofnitric oxide (NO) from a protein bonding site (heme or copper) found incytochrome c oxidase (CCO). CCO acts as a NO scavenger moleculeproviding negative feedback and NO regulation. As described earlier inreference to FIG. 2, in the presence of photobiomodulation, NO isreleased only where it is subjected to phototherapy, presumably only inthe locale of diseased or injured tissue. The observed benefit of pulsedphototherapy, it is postulated, occurs because pulsed light can triggermultiple photodissociation events, while in continuous wave (CW) modethe release of NO will stabilize at a lower fixed rate, balancing NOrelease with the counter-reaction of NO reattachment.

FIG. 13 schematically summarizes the physical mechanisms ofphotobiomodulation. As shown, photon 190 is absorbed by and interactswith molecule 191 to make or break new bonds. The energy of theimpinging light depends on its wavelength as given by the Einsteinrelation E=hc/λ, or for convenience sake by the relation E=1.24 eV-μm/λ,where λ, is measured in μm. For 650 nm red light E=1.91 eV per photonwhile for 950 nm NIR light E=1.31 eV. While most chemical bonds,including hydrogen, ionic, and most covalent bonds range in bondenergies from 0.2 eV to 10 eV, the making or breaking of a chemical bondfrom the energy of a photon is complicated by the fact that moleculesand especially crystals comprise groups of atoms with many bonds workingcollectively, meaning breaking a single bond does not necessarily inducea bond transformation. Moreover, depending on the reaction, multiplesources of energy and enzymes may assist the photon in inducing achemical transformation. For example, a single ATP molecule may releaseup to 0.6 eV of energy, thereby assisting singularly or collectively infueling a photochemical reaction.

The result of the photobiomodulation of molecule 191 may be manifestitself in one of several mechanisms, namely electrical conduction 192,chemical transformations 193, ionic conduction 194, or thermal vibration195. Release of free electrons 192 during ionization describes thepurely electrical component of photobiomodulation. Electron transportoccurring with a time constant τ_(e) is relatively fast and capable ofresponding to stimuli from kHz up to tens of kHz. Photobiomodulationinducing electrical conduction through electron emission and electrontransport can be referred to as biophotoelectric conduction.

Chemical transitions 193 along with ionic electrical conduction 194having respective time constants τ_(c) and τ_(Q) are slower, respondingto photobiomodulation in to 10 Hz to the 1 kHz range. Chemical processesare complex, involving a structural transformation in the affectedmolecule 198 with a corresponding change in its chemical reactivity andits stored potential energy (PE). Ionic processes 194 are significantlyslower than simple electron 192 conduction, because the conducting ions197 are oftentimes large molecules conducting by diffusion (driven by aconcentration gradient dN_(q)/dx) or by electrical conduction driven byintra- and intercellular electric field induced force qE, said electricfields existing as a result of spatially unevenly distributed ions.Photobiomodulation inducing electrical conduction through ionictransport can be referred to as biophotoionic conduction. Similarlyphotobiomodulation inducing structural transformations in molecules canbe referred to as biophotochemical transformation.

The other mechanism, thermal vibrations 195 is the spread of heat,either classical kinetic energy or by quantized phonon conductioncausing molecules 196 to vibrate at increased levels compared to theirsurroundings as energy escapes the photo-excited molecules and spreadsthermally into its neighbors. Transient thermal effects, vibrationsspreading across tissue can occur at a rate of 1 to 10 Hz while steadystate conduction can take minutes to stabilize, i.e. responding tosub-Hz frequencies. Thermal vibration is another important mechanism inphotobiomodulation because thermal excitation increases reaction ratesby causing interacting ions and molecules to bump into one another morefrequently and rapidly, the molecular version of stirring reactantchemicals in solution. Photobiomodulation inducing the diffusion of heatbetween and among molecules can be referred to as “biophotothermal”conduction or thermal vibration.

Frequency dependent photobiomodulation results from these physicalprocesses interacting with the modulating or pulse frequencies ofincoming photons. Overstimulation occurs when the digital pulse rate orlight modulating frequency is faster then the physical process's abilityto respond to it. In such cases, the response is reduced because thecell or molecule simply cannot keep up with the stimulus. Such a case isanalogous to a busy freeway with entrance ramp metering lights stuck-oncausing more-and-more cars to jam onto the freeway until no one is ableto move. Understimulation occurs when the digital pulse rate or lightmodulating frequency is much slower than the cell's ability to absorb itin which case little or no photobiomodulation occurs. This condition isanalogous to a freeway whose metering lights are allowing almost no oneto get onto the freeway, with the similar result that no one getsanywhere. Only if the photobiomodulation frequency matches the system'snatural response frequency is there an optimum result and efficientenergy transfer. For example, if the metering lights onto the freewayare timed correctly, the optimum number of cars will fill the freewayand promptly travel to their destination without starting a traffic jam.

As detailed, understimulation at too low of a frequency andoverstimulation at too high of a frequency results in a diminishedphotobiomodulation response, and only in between, at the optimum pulserate or excitation frequency can the photobiomodulation response andphototherapeutic efficacy be maximized. This peak response conditionoccurring at a particular frequency appears very similar to the resonantcurve of FIG. 10, especially since the prior analysis reveals multipletime constants exist in any cells, tissue or organs, each optimized toinduce specific electrical, ionic, chemical and thermal mechanisms.

Therefore, the various peak response conditions can be referred to asbioresonance even though the mechanism may not involve energy storageand timed release as in the true resonant systems described above. Beingable to stimulate these select resonant frequencies in a controlledmanner free from spectral contamination is critical, especially inavoiding the inadvertent generation of frequencies causing destructiveinterference and loss of efficacy. Moreover, invoking multiplebioresonant mechanisms simultaneously is not possible using present-daydigital pulse based phototherapy systems. The disclosed new electronicdrive system described herein comprises both an inventive apparatus andnovel methods for realizing sinusoidal drive and arbitrary waveformsynthesis of LED or laser light for phototherapy, not available or evensuggested in the prior art.

Waveform Synthesis System for Phototherapy

A key element in driving LEDs and laser diodes with controlledfrequencies and harmonics is the circuitry and algorithms used ingenerating the device's waveforms, patterns, and driving conditions.While the following description details the means to drive arrays ofmultiple strings of series-connected LEDs, the same circuitry can beadapted to drive one or multiple semiconductor lasers.

Because the light output of an LED depends on its current and becauseits brightness is poorly correlated to the forward voltage presentacross the LED during conduction, it is preferable to use controlledcurrent-sources (and current sinks) rather than constant voltage drive.For example if an series-connected string of LEDs is powered by avoltage source connected through a series resistor, the LED currentI_(LED) will unavoidably vary with the total series forward voltage dropV_(f) of all the LEDs. Provided the power supply voltage +V_(LED) ishigher than the forward voltage drop V_(f) of the LED string, i.e.+V_(LED)>V_(f), then the LED current I_(LED) is given by the seriesrelation I_(LED)=(+V_(LED)−V_(f))/R illustrating that any change in theLED voltage will result in changes in LED current and hence in LEDbrightness. Since LED voltages cannot be accurately controlled ormatched, unless each LED string comprises sorted LEDs with matched totalvoltages, any given LED string will invariably be brighter or dimmerthan the next.

FIG. 14 illustrates two equivalent representations 200 a and 200 b of acurrent sink controlling the current through a string ofseries-connected LEDs 205 a. In schematic 200 a, current sink 201 arepresents an idealized current controlled device with sensing andfeedback designed to maintain a prescribed current I_(LEDa) in LEDstring 205 a. As shown, the LED string 205 a comprises “m”anode-to-cathode series-connected LEDs, where m is a mathematicalvariable representing the number of LEDs in string 206. The schematicelement 199 a represents feedback from sensing the value of currentI_(LEDa) and using feedback to insure the current stays constant even ifthe voltage across current sink 201 a varies.

When conducting the value of the LED current I_(LEDa) is proportional toan analog input current αI_(ref) set by low-voltage current source 202a. When current sink 201 a is not conducting, i.e. when current source202 a is not enabled, the voltage across the LEDs is minimal and thevoltage supported across current sink 201 a approaches the value+V_(LED), a relatively high voltage, e.g. 40V, compared to the lowervoltage of +V_(logic), typically 3V to 5V. Current sink 201 a may bedigitally toggled on or off, i.e. conducting or not conducting, throughits digital enable pin labeled “Enable” connected to digital synthesizer203 a. Note that the letter “a” represents one of multiple channelsdriving separate series-connected strings of LEDs in an LED pad. LEDpads may contain many independently controlled strings of LEDs, namelyLED output channels a, b, c, . . . , n, where “n” is a mathematicalvariable representing the number of channels.

In schematic circuit 200 b, the series connection of “m” LEDs issymbolically replaced by a single LED with the number “m” inside thedevice and the voltage +V_(fθ) labeled across the LED. As shown, currentsink 201 a is further detailed showing an analog feedback circuitcomprising MOSFET driver 215 a driving the gate of high-voltage MOSFET216 a. In operation, MOSFET driver 215 a provides a voltage on the gateof current-sink MOSFET 216 a allowing current I_(LEDa) to flow throughthe sensing circuitry contained with MOSFET driver 215 a to ground. Thiscurrent is then compared to a multiple β_(r) of the analog input currentαI_(ref) set by low-voltage current source 202 a, and the gate voltageon current-sink MOSFET 216 a automatically adjusted by the circuitrywithin MOSFET driver 215 a until the multiple β_(r) of the currentαI_(ref) and the current I_(LEDa) match and I_(LEDa) is at is desiredvalue. Because of its analog closed-loop circuitry, feedback from MOSFETdriver 215 a is nearly instantaneous, adjusting dynamically withfluctuating voltages and programmed changes in the reference currentinput from current source 202 a.

The reference current αI_(ref) from current source 202 a may be realizedby a fixed, time varying, or adjustable reference voltage and a seriesresistor trimmed for accuracy to convert the precise voltage into aprecise reference current. The accurate voltage source may comprise afixed-value Zener diode or a bandgap voltage, a voltage-controlledoscillator (VCO), or a digital-to-analog converter (DAC) facilitatingdigital control of the analog current value output from current-source202 a. The digital pulse output from digital synthesizer 203 a can berealized by counters and clock circuits, by programmable logic arrays(PLAs), or by a microprocessor executing firmware or softwareinstructions.

Some implementations of the aforementioned circuitry are described in apreviously-cited related U.S. Pat. No. 9,877,361. Other exemplary andnovel analog, digital and mixed-mode circuits will be included hereinlater in the application.

FIG. 15 illustrates the diverse variety of waveforms that may besynthesized by the described driver circuitry. As shown, graph 240 aillustrates the input waveforms of current sink 201 a comprising thedigital Enable signal output from digital synthesizer 203 a, and thereference current αI_(ref) output from current source 202 a. Graph 240 billustrates the resulting LED current conduction waveform with the sametime references t₁, t₂, etc. as graph 240 a included for easycomparison. The generated waveforms are examples, not intended to implyany specific operating condition attempting to avoid undesirableharmonics in phototherapy systems, but simply to illustrate that thecombination of digital pulsing and analog current control offers nearlylimitless control of LED excitation.

As shown in graph 240 a, the digital Enable signal comprises linesegments 241 through 245, and reference current αI_(ref) comprisescurves 251 through 258. In the corresponding output of LED current ingraph 240 b, the instantaneous LED current is illustrated by curves 260through 269, while the average LED current, where applicable, isrepresented by the dashed lines shown by line segments 271 through 275.

To understand the interaction between the analog and digital control ofLED excitation, we will compare the two graphs in each correspondingtime interval. Specifically, before time t₁, enable signal 241 is at alogic zero and reference current 251 is biased at some nominal valueαI_(ref), e.g. at an input current corresponding to an I_(LEDa) outputcurrent of 20 mA. Because digital enable signal 241 is at a logic zero,the LED current 260 is at zero and the string of LEDs remains offdespite the non-zero value of reference current αI_(ref).

Between time t₁ and t₂ digital enable signal 242 jumps from a logic zerostate to a logic one state while the value of reference current 251remains biased at a value of αI_(ref) for example at 20 mA to 30 mA. Asa result, LED current 261 jumps to the value of reference current 251.The off-to-on transition in LED conduction at time t₁ illustrates theeffect of digitally “toggling” an analog current sink.

While digital enable signal 242 remains on, at time t₂ the analogmagnitude αI_(ref) of reference current 252 jumps to a higher value andthen declines in a specific but user settable manner until it finallysettles at a value 253, which is the same as its original value 251. TheLED current 262 similarly tracks the reference, jumping from 20 mA to ahigher value, e.g. 27 mA, before settling back at 20 mA at time t₃,shown by LED current 263. The output waveform of LED currents 262 and263 illustrates that the reference current can be used to facilitatepurely analog control of LED current and brightness with no digitalpulsing whatsoever.

At time t₄, as shown by curve 254, the reference current commences acontrolled, small signal sinusoidal oscillation superimposed on annon-zero average DC value. The perturbation in the reference current maybe considered small-signal because the amplitude of the oscillation issmall compared to the average value of current αI_(ref). As a symmetricoscillation, the average current remains unchanged from the DC value(shown by curve 253) of the reference current existing before theoscillations commenced. While any oscillating frequency may beconsidered possible, practical considerations and the value ofoscillating waveforms in phototherapy suggest the operating frequencyshould be 20 kHz or below. The corresponding LED current, depicted ascurve 264 in graph 240 b commencing at time t₄, tracks that of thereference current shown by curve 254, having an average current value(dashed line 271) of 20 mA and varying symmetrically around the averageLED current by some fixed amount, for example ±1 mA. This means that theLED current varies sinusoidally, with peak-to-peak values ranging from19 mA to 21 mA.

At time t₅, as shown by curve 255, the small signal oscillations in thereference current during the prior interval t₄-t₅ grow into large signaloscillations shown by curve 255 and having the same frequency ofoscillation as the prior interval. In the example shown the minimumreference current αI_(ref) reaches zero (or nearly so) while the peakreference current reaches twice the average value, i.e. twice the valueof the reference current represented by curve 253. As before, since thevalue of the digital enable signal (line segment 242) remains at a logicone state, the LED current (curve 265) tracks the value as a multiple ofthe reference current (curve 255) both in frequency and in wave shape,having an average LED current (dashed line 271) of 20 mA withpeak-to-peak oscillations around that average of nearly ±20 mA, meaningthe LED current varies sinusoidally from 0 mA to 40 mA with an averagevalue of 20 mA.

Starting at time t₆, the same oscillatory operating conditions persistas existed in the interval t₅-t₆, except that the oscillation frequencyof the reference current represented by curve 255 and correspond LEDcurrent represented by curve 265 is intentionally reduced to a loweroscillating frequency, shown by curve 256 for the reference current andby curve 266 for the corresponding LED current, with the output stillmaintaining an average LED current 71 of 20 mA, the same average aspreviously occurred for oscillatory LED currents shown by curves 264 and265.

At time t₇, the roles of the digital enable signal and the referencecurrent αI_(ref) are reversed, whereby the value of reference currentbecomes constant at some nominal value (shown by line segment 257) andthe digital enable signal begins pulsed operation. Specifically at timet₇, the digital enable signal (shown by curve 243) commences pulsedoperation with 50% duty factor, pulsing at a digital clock frequency of1/T₁, where T₁ is the period of each repeated cycle. At time t₈, asshown by curve 244, the pulse on-time of digital enable signal increaseswhile the period T₁ and the corresponding pulse frequency remain thesame as before. As a result, the 20 mA pulses of LED current at a 50%duty factor, represented by curve 267, become an LED current become anLED current at a 75% duty factor, represented by curve 268. This mode ofoperation comprises fixed-frequency PWM or pulsed width modulationoperation, where the average LED current varies from 50% of 20 mA, i.e.10 mA average LED current (represented by dashed line 272), to 75% of 20mA or 15 mA average LED current (represented by dashed line 273) at timet₈.

At time t₉, while the value of the reference current remains unchanged(curve 257), the period of the pulses of the digital enable signalincreases to a value T₂, as does the pulse on time, as shown by curve245. This is reflected by the waveform of the LED current (curve 269).As shown, the duty factor, the pulse on time of the digital enablesignal represented by curve 245, divided by the total period T₂ alsoincreases, resulting in the LED current having a higher average value(shown by dashed line 274), corresponding to an increase of duty factorto 90%. The reduction in operating frequency from 1/T₁ during theinterval between times t₇ to t₉, to the lower operating frequency 1/T₂thereafter is an example of variable frequency PWM operation, andclarifies that PWM duty factor can be varied independently of thedigital pulse frequency.

In the final waveform shown in FIG. 15, at time t₁₀ the value ofreference current increases to a higher value (represented by thetransition from curve 257 to curve 258), while the waveform of digitalenable signal remains the same as it was in the prior interval t₉-t₁₀.The result is that the instantaneous value of the LED output currentincreases, as shown by the transition from curve 269 to curve 270 andthe average LED current also increases, as shown by the transition fromthe dashed line 274 to the dashed line 275. Despite increasing theaverage and instantaneous LED brightness, the duty factor and the pulsefrequency of the LED current remain unchanged from the correspondingvalues in the time interval t₉-t₁₀.

In conclusion, the instantaneous and time average value of the LEDcurrent can be controlled in numerous and flexible ways using analogcontrol of the reference current and digital pulse control of the enablesignal of the current sink schematic representations shown in FIG. 14.Realizing current sink 215 a, reference current source 202 a, anddigital synthesizer 203 a can be accomplished in many ways. Actualrealization of these circuits must address issues of accuracy,reproducibility, and scalability into multichannel systems. Suchcircuitry can be divided into two broad categories—analog LED controland digital synthesis.

Analog LED Current Control

Referring again to FIG. 14, controlling LED current I_(LEDa) requiresanalog control to implement the sense and LED drive circuitry withinMOSFET driver 215 a, as well as to implement precision reference currentI_(ref).

Current sink 201 a comprises high-voltage MOSFET 216 a biased to controlthe LED current I_(LEDa) and MOSFET driver 215 a which senses the LEDcurrent I_(LEDa) compares the LED current I_(LEDa) to the desiredreference current αI_(ref) and dynamically adjusts the gate voltage onhigh-voltage MOSFET 216 a until the LED current I_(LEDa) matches thepredefined scalar multiple β_(r) of the reference current αI_(ref).Measurement and feedback must operate in a closed loop manner to adjustfor any manufacturing variations in high-voltage MOSFET 216 a affectingits transconductance and channel-to-channel matching such as thresholdvoltage and gate oxide thickness.

Although the reference current αI_(ref) is schematically represented asa controlled current, distributing precise currents across multiplechannels, as shown in FIG. 16A, is problematic because the total currentnαIref from current source 206 will not necessarily be distributedevenly among the inputs to MOSFET drivers 215 a through 215 n, i.e.I_(a)≠I_(b)≠I_(n). The solution to this problem, shown conceptually inFIG. 16B, is to employ a reference voltage source 207 to distribute avoltage V_(ref) rather than a current to each channel and to convertthis voltage into identical currents using a transconductance amplifier208 a, 208 b . . . 208 n in each channel. For example, transconductanceamplifier 208 a converts V_(ref) into current I_(a) feeding MOSFETdriver 215 a, transconductance amplifier 208 b converts the same V_(ref)into current I_(b) feeding MOSFET driver 215 b, and so on.

In practice however, it is unnecessary to employ n-channels oftransconductance amplifiers since the voltage conversion function can beperformed inside the MOSFET driver's circuitry. For example, as shown inFIG. 16C, the current I_(a) coming from reference voltage source 207 andfeeding MOSFET driver 215 a is used to bias a current mirror MOSFET 210through bias resistor 212 and a parallel network 220 of trim resistors213 a through 213 x. The subscript “x” is a mathematical variablerepresenting the number of resistors in parallel network 220. Since thegate of MOSFET 210 is connected to its drain, i.e. MOSFET 210 is“threshold connected,” the gate voltage of MOSFET 210 will naturallybias itself to a voltage V_(G2) sufficient to conduct the desiredreference current I_(a) as set by series resistor 212 and the paralleltrim network 220 comprising resistors 213 a through 213 x. MOSFET 210and the parallel combination of resistor 212 and trim network 220 form avoltage divider, where the voltage across mirror MOSFET 210,V_(pilot)=V_(ref)−I_(a)·R_(equiv) where1/R_(equiv)=1/R_(max)+1/R_(t1)+1/R_(t2)+ . . . +1/R_(tx). By changingthe resistive value of the resistor network 220, V_(pilot) adjustsitself to produce a gate voltage V_(G2) on MOSFET 210 consistent withits drain current because its gate and drain are connected, i.e.V_(GS)=V_(pilot). The gate voltage V_(G2) of MOSFET 210 will be slightlylarger than its threshold voltage, hence the designation “thresholdconnected.”

This same gate voltage V_(G2) biases a much larger MOSFET 211 to thesame gate drive condition such that the ratio of nominal operatingcurrents through current mirror MOSFETs 210 and 211 (i.e., theabove-defined multiple β_(r)) is equal to the ratio of the gate widthsof current mirror MOSFETs 210 and 211. For example, if reference currentI_(a) is nominally set at 2 μA and I_(LEDa) is intended to be 20 mA,then the size ratio between MOSFETs 210 and 211 should be selected to be20 mA/2 μA=10,000, meaning that the gate width of current mirror MOSFET211 should be 10,000 times larger than the gate width of MOSFET 210.Because of their common gate biasing and fixed size ratio, only whencurrent mirror MOSFET 211 is conducting 20 mA, will its drain-to-sourcevoltage V_(sense) be equal to V_(pilot). During the illumination of LEDstring 205 a, a differential amplifier 214, which is biased in a closedloop with a stable voltage gain Av, drives the gate of high-voltageMOSFET 216 a with a gate voltage V_(G1), till the current I_(LEDa)flowing in MOSFETs 216 a and 211 drives the difference between V_(sense)and V_(pilot) to zero, i.e. V_(sense)−V_(pilot)=0. In this way, themultiple β_(r) of the reference current I_(a) is “mirrored” in MOSFET211, and a controlled and constant current flows in LED string 205 aeven if the LED supply voltage +V_(LED) changes.

During manufacturing, the resistor network 220 in parallel with fixedresistor 212 is functionally trimmed to produce an accurate outputcurrent thereby eliminating the impact of variability coming from MOSFETtransconductance of MOSFET 210 or in the resistor value R_(max) ofresistor 212. In the example shown, trimming is performed by measuringthe current I_(LEDa) and then blowing fuse links till the measured valueof I_(LEDa) reaches its target value. Because amplifier 214 controls thegate voltage of MOSFET 216 a (and hence the current I_(LEDa)) andprovided the size of MOSFETs 210 and 211 are equal, then the errorvoltage, the difference between V_(sense) and V_(pilot), will be drivento zero when the currents I_(a) and I_(LEDa) are equal. Should the gatewidth of MOSFET 211 be larger than that of MOSFET 210, then when theerror voltage is zero, the LED current I_(LEDa) will be larger thanreference current I_(a) by the ratio between the respective gate widthsof MOSFETs 211 and 210, which as indicated above is equal to β_(r).

For example, initially after manufacturing and immediately prior totrimming when all the resistors in resistor network 220 are stillelectrically connected in parallel with resistor 212, the totalresistance of the resistor network 212 is at its minimum value, I_(a) ishigher than its target value, and therefore the value of I_(LEDa) willalso be too high, e.g. 22 mA (10% above its target value of 20 mA). Onthe integrated circuit (or on a printed circuit board) probes areelectrically connected to common metal trim pad 221 and to all thespecific resistor trim pads 222. For clarity's sake, only trim pad 222 bin series with trim resistor 213 b is labeled. A high current is thenimpressed by the tester between common trim pad 211 and a specificchannel's trim pad, e.g. trim pad 222 b, causing the thin portion of themetal fuse link 223 b in series with trim resistor 213 b to melt andbecome an electrical open circuit, disconnecting resistor 213 b fromtrim network 220. With less parallel resistance, the total resistanceincreases, the value of reference current drops, and the LED current inLED string 205 a decreases by a fixed amount.

This measurement and link blowing process is repeated until the propernumber of metal fuse links have been blown reduce the magnitude of thereference current I_(ref) by the factor α so as to produce the targetvalue of the current I_(LEDa). αI_(ref) thus represents the referencecurrent after trimming. If all the fuse links are blown, the resistancein series with MOSFET 210 increases to its maximum value R_(max), theresistance of resistor 212, and reference current I_(a) reaches itslowest value. If that current is still above the target value, then thatparticular integrated circuit will be rejected as defective, loweringproduction yield and increasing product cost. As such, the resistancevalues R_(t1), R_(t2), . . . R_(tx) used in resistor network 220 must bechosen carefully to accommodate normal stochastic variability inintegrated circuit manufacturing. Note that the schematic representationof fuse link 223 b is illustrated by a line that is thinner than therest of the conductors shown in the schematic of FIG. 16C.

Also, single-pole double-throw switch 217 is shown to illustrate thedigital enable function within MOSFET driver 215 a. When the digitalinput to digital gate buffer (shown as an inverter symbol) 218 is “high”or a logic one, switch 217 connects the gate of high-voltage MOSFET 216a to the output terminal of differential amplifier 214, turning MOSFET216 a on and illuminating LED string 205 a. If the enable signal isbiased to a logic zero state, the switch 217 connects the gate ofhigh-voltage MOSFET 216 a to ground, whereby V_(G1)=0 and MOSFET 216 aturns off, cutting off the current in LED string 205 a. While thisfunction is shown as a mechanical switch, it is actually realized by anetwork of transistors configured as an analog switch or amplifier ascommonly known to those skilled in the art. Also, during times when aspecific channel is not enabled, the operation of differential amplifier214 may be suspended or clamped in voltage so that it does not try toincrease its output voltage in a futile attempt to increase the sensecurrent in MOSFET 211.

While resistor trimming is commonplace, trimming the size, i.e. gatewidth, of a network of transistors is generally easier and more accurateand reproducible than using resistors. Such a circuit is shown in FIG.16D, where resistor 212 has no parallel network of trim resistors butinstead current mirror MOSFET 210 includes a parallel network 230 oftrim MOSFETs 225 a, 225 b . . . 225 x. Another advantage of using MOSFETtrimming rather than resistor trimming is that network 230 is generallysmaller than network 220, shown in FIG. 16C. Like the resistor trimmethod, as shown fuse links (illustrated by fuse link 233 x) are blownto disconnect, i.e. turn off, one or more of MOSFETs 225 a . . . 225X inparallel with current mirror MOSFET 210. For example, initially aftermanufacturing and immediately prior to trimming, when all of the MOSFETs210 and 225 a . . . 225 x are still connected in parallel, the sizeratio between MOSFET 211 and the parallel combination of current mirrorMOSFET 210 and trim network 230 is at a minimum and the current I_(LED)will be below its targeted value, e.g. at 18 mA, 10% below its 20 mAtarget. By forcing a high current between common trim pad 231 andchannel specific trim pad 232 x, for example, fuse link 233 x is blownand the gate of trim MOSFET 225 x is no longer connected to the gate ofMOSFET 210. Instead, with its gate disconnected, resistor 226 x biasesMOSFET 225 x off. With less parallel gate width in MOSFET trim network230, the current mirror gate ratio increases and for the same value ofreference current I_(a), the LED current I_(LED) will increasecommensurately.

Note that, in FIG. 16D, the gates of MOSFETs 210 and 211 along withthose in MOSFET trim network 230 are biased by a voltage source 224 andnot by connecting the gate of current mirror MOSFET 210 to its drain.The advantage of this method is that the current mirror MOSFET 211 mayoperate with a lower drain voltage V_(sense) using this method. Whilesome initial accuracy may be lost using this method, functional trimmingis able to correct for this deficiency. Beneficially, the lower voltagedrop across MOSFET 211 reduces power dissipation and improves overallsystem efficiency of the LED driver 215 a.

An alternative method of trimming involves varying the reference currentby adjusting the reference voltage. Adjusting the reference voltage alsorequires analog circuitry. Methods of manufacturing fixed valuereference voltage sources are well known, including means to minimizevariation in the voltage over temperature. Such methods include bandgapvoltage references (see en.wikipedia.org/wiki/bandgap_voltage_reference)and Zener diode voltage references (seeen.wikipedia.org/wiki/Zener_diode). Since these techniques are wellknown to those skilled in the art, they will not be discussed here.

Analog Sinusoidal Synthesis

While sinusoidal waveforms can be generated digitally as described laterin this application, an inventive means disclosed herein by which tosynthesize a sinusoidal waveform for driving LEDs in a phototherapysystem is through the use of analog synthesis. While digital synthesis,as disclosed, involves pulsing an LED current on-and-off in constantlyvarying durations, i.e. pulse-width-modulation, to synthesize a sinewave (or chords of multiple frequency sine waves), analog synthesisinvolves sinusoidally varying the reference current or current bias tothe LED current control circuit, i.e. the current mirror driving an LEDstring, in essence making the reference current into an oscillator.Referring to the exemplary waveforms shown in FIG. 15, analog waveformsynthesis is illustrated by sinusoids 254, 255 and 256 occurring attimes t₄, t₅, and t₆, and also by the arbitrary time dependent waveformrepresenting the ability to implement any control function by waveform252 at time t₂.

As shown in FIG. 17A, to perform analog sinusoidal synthesis, thereference voltage biasing MOSFET driver 215 a is replaced with a fixedfrequency sine-wave or sinusoidal oscillating reference voltage source235, also known as a linear or “harmonic” oscillator. Harmonicoscillators in the audio range can be made using inductor-capacitor,i.e. LC, oscillators or using resistor-capacitor, i.e. RC, oscillatorscircuits including RC phase shift oscillators, Wien bridge oscillators,or twin-T oscillators (see wikieducator.org/sinusoidal_oscillator).During manufacturing, the output voltage of oscillating referencevoltage source 235 must be trimmed using resistors or transistor arraysin a manner similar to the trimming of MOSFET driver 215 a describedpreviously. In contrast, other common RC circuits often used for clockgeneration comprising simple relaxation oscillators are not harmonicoscillators and are not applicable because they produce sawtooth ortriangular shaped waveforms with unwanted broadband spectral content.

In FIG. 17B, oscillating reference voltage source 235 is replaced by acontrolled oscillating reference voltage source 236 with an adjustablefrequency and an adjustable voltage. One example of such an oscillatingreference is illustrated in FIG. 17C comprising a Wien oscillator 280with a voltage follower 281 and a trimable variable voltage outputbuffer 282. Wien oscillator 280 comprises two matched variablecapacitors 284 a and 284 b and two matched programmable resistors 283 aand 283 b. The two RC networks create a voltage divider and feedbacknetwork returning signals from the output of a high-gain differentialamplifier 285 back to its positive input. A damping network comprisingresistors 286 a and 286 b sets the gain and stability of the circuit tostabilize the oscillations.

The oscillating frequency may be adjusted by changing the resistanceR_(osc) of programmable resistors 283 a and 283 b or alternatively bychanging the capacitance C_(osc) of variable capacitors 284 a and 284 b.Variable resistance may be realized by varying the gate voltage andresistance of MOSFETs biased in their linear region of operation, oralternatively using a digital potentiometer comprising discreteresistors with parallel MOSFETs able to short out the various resistors.Variable capacitance may be realized by varactors comprisingback-to-back PN junction diodes, one of which is reverse biased to afixed voltage to establish the junction capacitance. Changing either theresistance or the capacitance adjusts the oscillating frequency of Wienoscillator 280.

To insure that loading by trimable variable voltage output buffer 282does not affect the oscillating frequency of Wien oscillator 280,voltage follower 281 comprising a differential amplifier 287 withnegative feedback through resistor 288 provides buffering. The voltageV_(buf) of voltage follower 281 is then adjusted by a resistor dividercomprising a fixed resistor 292 and a variable resistor 291 withresistance values R₁ and R₂ respectively. The variable resistance 291may comprise a trim network as well as a digital potentiometer, asdescribed previously. The voltage at the tap point located betweenresistors 291 and 292 and connected to the positive input ofdifferential amplifier 289, is equal to the output voltage V_(ref)out ofvariable voltage output buffer 282 and is given byV_(ref)out=(V_(buf)·R₂)/(R₁+R₂). With its output connected to itsnegative input by wire 290, differential amplifier 289 behaves as avoltage follower faithfully reproducing the voltage waveform of itsinput while delivering the required current to an electrical loadconnected to its output V_(ref)out.

As shown by the output waveform 295, this output voltage V_(ref)out hasan AC component V_(AC)(t) extending from zero to its peak value of+V_(AC)(t) with an average value of V_(AC)(t)/2 and contains no added DCoffset (aside from the intrinsic DC average value of a sine wave). Sincethe only voltage component is AC, specifically the sine wave generatedfrom Wien oscillator 280+V_(AC)(t), then the sine wave can be said torepresent large-signal AC behavior. If it is desirable to also include aDC offset, the output of oscillating reference voltage source 236 may befurther adjusted by the circuit shown in FIG. 17D. In this circuit, theV_(ref)out output of the circuit shown in FIG. 17C is fed into a voltagefollower 300 comprising a differential amplifier 302 (or another type ofvoltage follower circuit) through an AC coupling capacitor 303.Differential amplifier 302 operates as a voltage follower because ofnegative feedback on wire 301, connecting its output to its negativeinput. The purpose of AC coupling capacitor 303 is to block any DCoffsets present within the output of oscillating reference voltagesource 236. If no offset is present capacitor 303 may be eliminated.

Although operational amplifier 302 is powered from logicsupply+V_(logic), its negative supply rail is not connected to groundbut instead is connected to a generated voltage +V_(neg) produced by avoltage bias circuit 309, an above ground voltage that acts as thenegative supply rail for differential amplifier 302. Because of thisre-referencing its negative supply rail, the output voltage V_(ref)out₂of differential amplifier 302 is shifted in its voltage level fromground to a more positive voltage. As a result, the waveform of theoutput voltage V_(ref)out₂ appears the same as the waveform of its inputV_(ref)out but V_(ref)out₂ is offset by a DC voltage equal to thegenerated voltage +V_(neg), or mathematically asV _(ref)out₂ =V _(DC) +V _(AC)(t)+V _(neg) +V _(ref)out₂ <+V _(logic)

The circuit will faithfully reproduce the input so long that the sum ofthe DC bias (+V_(neg)) and the sine wave input signal AC(t) do notexceed the supply voltage +V_(logic), otherwise the top of the sine wavewill be “clipped”, i.e. reach a constant maximum output voltage at+V_(logic) during any interval where +V_(neg)+V_(ref)out₂≥+V_(logic).Waveform clipping results in the distorting of the output waveform,producing unwanted harmonics and spectral contamination similar to (oreven worse than) that of LED drive using digital pulses. Also note thatif the difference in voltage (+V_(logic)−+V_(neg)) is too small, meaningthat the level shifted bias is too high, differential amplifier 302 maynot be able to function properly.

Generation of the DC voltage +V_(neg) may be performed in any number ofways including a trimmed bandgap voltage followed by a variable gainamplifier, a voltage controlled amplifier, or varying resistor orswitched-capacitor voltage divider networks. One such voltage dividermethod is illustrated in FIG. 17D as voltage generation circuit 309using a resistor voltage divider technique. As shown, the logic supplyvoltage +V_(logic) is connected to series resistor string comprisingresistors 304 a through 304 x, where x is a mathematical variablerepresenting the number of resistors in the series resistor string.Resistors 304 b through 304 x are connected in parallel with MOSFETs 305b through 305 x, respectively. The number of resistors may commonly be9, 13, or 17 allowing various 8-bit, 12,-bit and 16-bit combinations ofvoltage to be realized depending on the accuracy required, where thenumber of resistors needed equal one plus the number of bits of accuracydesired. For example, 8 bits of accuracy requires 9 resistors providing256 levels of output voltage.

The output voltage +V_(neg) taken from the voltage tap point betweenresistors 304 a and 304 b, is varied by shorting out various resistorsby turning on and off MOSFETs 305 b through 305 x in variouscombinations. For example, if all the MOSFETs 305 b through 305 x areturned on and their resistance is designed to be small relative to thatof the resistance value R₁ of resistor 304 a, then the output voltage+V_(neg) is near ground; if none of the transistors 305 b through 305 xis turned on, the output voltage +V_(neg) becomes +V_(logic), and forvarious other combinations an intermediate voltage may be selected. Theresistor network 304 a through 304 x can further be modified to select avoltage from only a portion of the supply range. For example, a lowervoltage than +V_(logic) may be used to power the resistor string. Theseries ladder of resistors 304 a through 304 x forms a type ofdigital-to-analog converter because turning various MOSFETs on and offis essentially a digital function and the result is an analog, albeitquantized, voltage. For greater resolution the number of resistors canbe increased or the voltage range reduced to that the least significantbit, i.e. LSB, represents a smaller voltage gradation.

Aside from the voltage generator function, resistor trim network 310comprising parallel resistors 311 a through 311 x is placed in parallelwith resistor 304 a to provide a means to trim the voltage accuracyduring manufacturing by blowing fuse links by impressing temporary highcurrents on trim pads on an IC. For example, by running high currentbetween common trim pad 312 and trim pad 314, thin metal line 313 willact like a fuse and melt, creating an electrical open-circuit andremoving resistor 311 b from the parallel network of resistors in trimnetwork 310.

In conclusion, the DC offset circuit shown in FIG. 17D, combined withthe oscillating reference voltage circuit of FIG. 17C allow theelectrical generation of a sine wave AC(t) of varying frequency andmagnitude offset by a DC voltage. So long as it does not exceed thesupply voltage +V_(logic), the output voltage of this newly disclosedoscillating reference voltage isV_(ref)out₂=V_(DC)±V_(AC)(t)/2=+V_(neg)±V_(ref)out₂ having a peak outputvoltage of V_(DC)+V_(AC)(t)/2, a minimum output voltage ofV_(DC)−V_(AC)(t)/2, and an average output voltage of V_(DC). If the ACcoupling capacitor 303 is removed, the average value of the outputincreases by the average voltage of the sign wave V_(AC)(t)/2, reducingthe usable operating voltage range of differential amplifier 302.

As shown by the V_(ref)out₂ waveform 308, using the circuit of FIG. 17Dor a similar circuit, the AC component of the signal is smaller than theDC offset voltage, i.e. V_(AC)(t)<V_(DC). Since the main voltagecomponent is DC and not the sinusoid, then the sine wave can be said torepresent small-signal AC behavior. In a phototherapy application, thevoltage value of V_(ref)out₂ actually represents the reference currentthat determines LED brightness whenever the LED string is enabled andconducting. Small signal operation of the inventive circuitry representsa completely new operating mode for phototherapy—one wherein the LEDstring is continuously illuminated at a fixed current and then modulatedsinusoidally at bias condition with slight increases and decreases incurrent and corresponding changes in brightness.

As shown in FIG. 18A, another way to vary the reference current is tosupply the reference voltage used to generate the reference currentαI_(ref) for MOSFET driver 215 a from a digital-to-analog (D/A)converter 315. While any number of bits may be used to control accuracy,commonly available converters, for example those used in HDTVs, comprise8 bits with 256 levels, 12 bits with 4096 levels, or 16 bits with 65,536levels. Converter speed is not high because the highest frequencyrequired for phototherapy is 20 kHz, and in most cases only 5 kHz. Inoperation, data is written into a latch or static memory, specificallyI_(LED) register 316, and loaded into D/A converter each time theconverter receives a digital clock pulse on its Load input pin, i.e.between 5 kHz to 20 kHz, as desired.

While many methods including switched capacitor, resistor ladder andother types of D/A converters (DAC) exist, because only audiofrequencies are required in phototherapy applications low cost solutionsmay be utilized. One such circuit is an 8-bit resistor ladder converter315 shown in FIG. 18B comprising a precision reference voltage source320, and a DAC resistor ladder comprising resistors 321 a through 321 x,along with DAC switches comprising MOSFETs 322 b through 322 xcontrolled by decoder 323. MOSFETs 322 b through 322 x are connected inparallel with resistors 321 b through 321 x, respectively. In operation,a decoder 323 loads an 8-bit word from its input line 8 b upon receivinga clock pulse on its digital Load input, represented by digital inverter344, and converts the 8-bit word into instructions of which of theMOSFETs 322 b through 322 x should be turned-on in various combinationsto produce a linear output voltage on the DAC ladder tap point betweenresistors 321 a and 321 b. The DAC ladder voltage, ranging from zero toV_(ref), is then fed to the positive input of a differential amplifier335 configured as a voltage follower. A resistor trim network 325comprising resistors 324 a through 324 x, trim pads (e.g. 326 and 328)and fuse links 327, is placed in parallel with resistor 321 a in orderto trim the output voltage during manufacturing. Alternatively, theinternal reference voltage V_(ref) provided by source 320 may be trimmedto provide the required precision.

As an inventive element, a switched filter capacitor 342 is optionallyincluded to filter the ripple of the output voltage V_(ref)out, or if ahigh speed transient is desired to disable the filter depending on thedigital control signal on the Filter Enable input represented by digitalinverter 343. In operation when MOSFET 340 is turned on and MOSFET 341is disabled capacitor 342 is connected in parallel with the output ofbuffer amplifier 335 and the output of reference 315 is filteredremoving high frequency noise. When MOSFET 340 is turned off and MOSFET341 is enabled, capacitor 342 is disconnected from the output of bufferamplifier 335 and the output of reference 315 is not filtered. Byenabling MOSFET 341, the charge on capacitor 342 is discharged toprevent the accumulation of voltage from repeated operation. Other D/Aconverters may be employed in place of resistor ladder converter 315, asdesired.

An example of an 292 Hz (D4) oscillating reference voltage without anyadded DC offset generated in the disclosed manner is illustrated in FIG.19A comprising a 1.2V sine wave 371 with a period of 3.42 msec and anaverage voltage output of 0.6V. The peak voltage is conveniently chosento be similar to the output voltage of a bandgap voltage trimmed for alow temperature coefficient or near zero “tempco”. Other voltages,however, may be employed as well to produce the desired input current toLED driver 215 a.

It should be emphasized that sinusoid 350 as disclosed herein issynthesized, programmable, and low voltage, not the artifact of arotating electromagnetic generator or alternator used in AC powergeneration in power plants. So while LEDs used in residential andcommercial lighting applications can, at least theoretically, be drivendirectly from the 60 Hz AC line voltage, the sinusoidal characteristicof the AC line voltages and its application in general lighting iscompletely different than the proposed synthesized sine wave excitationof LEDs applicable for phototherapy.

First, the AC line voltage is high-voltage, typically 110 VAC or 220 VACand unacceptably dangerous in medical applications where a device, inthis case the LED array and pad, touches the skin. In LED drive forphototherapy, the total number of series connected LEDs is limited tooperate at a maximum voltage below 40V, a voltage considered safe byUnderwriter Laboratories (UL) for consumer and medical applications.

Second, the frequency of the AC line varies with loading of the utilitycustomers and is contaminated by numerous undesirable spectral harmonicsaffecting the purity of the sinusoid and rendering it unsuitable forphototherapy applications.

Third, the frequency of the AC line, namely 60 Hz and its harmonic 120Hz do not represent a frequency known to be beneficial in phototherapy,e.g. a multiple of 292 Hz. In fact 60 Hz does not represent a multipleof any pure or chromatic tone indicated for photobiomodulation.

Fourth, aside from its uncontrolled variation with loading, thefrequency of the AC line is fixed and is not programmable or adjustable.It cannot be adjusted or varied dynamically or to match the timeconstants of natural biological processes and associated time constants.It also cannot be used to generate chords of multiple frequencysinusoids nor control the energy density and spectral content, i.e. themix, of multiple frequency sinusoids.

Fifth, the reduction of the AC mains line voltage from 110 VAC or 220VAC to a safe level, i.e. below 40V, requires a large and heavyiron-core transformer designed to operate at 60 Hz.

Sixth, LEDs used in phototherapy necessarily comprise relatively narrowspectral wavelengths in the red, near infrared, or blue portion of thespectrum. The LED light, typically ±35 nm in spectral width, emittedthrough the quantum-mechanical process of tunnel emission is determinedby bandgap engineering of the manmade crystal used to realize the LED inmanufacturing. LEDs used in lighting are designed to emit a broadspectrum of light, i.e. white light, comprising a number of colors inthe rainbow. Unlike LEDs used in phototherapy, white light LEDs compriseblue or UV LEDs with a lens cap containing phosphor tuned to absorb blueor UV light. In operation, the light emitted from the LED semiconductormaterial is absorbed by the phosphor atoms in the lens cap and convertedinto broad spectrum “white” light similar to sunlight but more white andless yellow.

Finally, the direct drive of LEDs using AC sinusoids in general lightingapplications is actually not in commercial practice today for a varietyof intractable technical problems including poor power efficiency, poorpower factor, electrical shock risk, and flicker. Today's LED bulbs usemultistage PWM switching power supplies for power factor correction andvoltage regulation. LED brightness is therefore controlled by digitalpulses and not using sinusoids.

So LEDs driven in AC lighting are not applicable for phototherapy.

In the operation of D/A converter 315, the digital input to decoder 323is repeatedly loaded during clocking of the Load pin, i.e. the input toinverter 344, occurring at fixed time intervals in order to generate asine wave of an arbitrary and adjustable frequency. The following tablerepresents examples of various time points used in the waveformsynthesis.

time DAC arc (msec) binary in hex level degree output V 0 0000 0000 0 0° 0.600000000 0.014 0000 0001 01 1   1.5° 0.615706169 0.029 0000 001002 2   3.0° 0.631401574 0.043 0000 0011 03 3   3.5° 0.647075457 0.1430000 1010 0A 10  15° 0.755291427 0.285 0001 0100 14 20  30° 0.9000000000.428 0001 1110 1E 30  45° 1.024264069 0.571 0010 1000 28 40  60°1.119615242 0.714 0011 0010 32 50  75° 1.179555496 0.856 0011 1100 3C 60 90° 1.200000000 1.284 0101 1010 5A 90 135° 1.024264069 1.427 0110 010064 100 150° 0.900000000 1.698 0111 0111 77 119  178.5° 0.615706169 1.7120111 1000 78 120 180° 0.600000000 1.727 0111 1001 79 121  181.5°0.584293831 1.998 1000 1100 8C 140 210° 0.300000000 2.569 1011 0100 B4180 270° 0.000000000 3.139 1101 1100 DC 220 330° 0.300000000 3.396 11101110 EE 238 357° 0.568598426 3.410 1110 1111 EF 239  358.5° 0.5842938313.425 1111 0000 F0 240 360° (0°) 0.600000000 skipped 1111 0001 F1 241not used N/A skipped 1111 1111 FF 255 not used N/A

As shown, an 8-bit D/A converter exhibits 256 output states or 256 stepsabove its zero state, i.e. from 0000-0000 in binary or from 00 to FF inhexadecimal. To conveniently map these states to the 360 degrees ofangle arc, only 240 steps (i.e. 241 states) of the D/A converter havebeen employed. As such, 240 steps corresponds to 360° or 1.5° per DACstep. The remaining DAC steps from 241 to 255, in hexadecimalcorresponding to DAC input codes from F0 to FF are intentionally skippedand not used in sinusoid generation. As described, the DAC value isrepresented in three equivalent ways

-   -   by a hexadecimal digital code, the input to decoder 323 in FIG.        18B, as illustrated by the hex code in the third column of the        above table    -   by a binary digital code shown in the second column of the above        table representing the various combinations of turning on and        off the MOSFETSs 322 b through 322 x in FIG. 18B to dynamically        change the resistor divider network ratio    -   by the analog output voltage output from DAC 315 and buffer 335        shown by the rightmost column in the above table, or        alternatively by a current in case that voltage is divided by a        resistor to make a DAC controlled current.

In operation, a sequence of increasing digital codes is fed into to theDAC at a regular time intervals to produce a rising output voltage.Conversely a sequence of declining digital codes may be used to lowerthe output voltage of the DAC. If this increasing and decreasing codesequence is performed repeatedly and consistently a any periodicfunction can be synthesized as an output of DAC 315. If codes are inputinto the DAC at regular time intervals according to evaluation of a sinefunction for fixed steps of angles, e.g. 15°, then the sequence willresult in a sinusoidal output from DAC 315.

To synthesize a 292 Hz sine wave having a period of approximately T=3.42msec, each of the 240 steps comprises 0.0142694 msec. The minimumcorresponding signal used to load DAC's decoder 323 must therefore be292 Hz·240 states/Hz or 70,080 Hz. The resulting spectra of theoscillating reference voltage is illustrated in FIG. 19B using a D/Aconverter to synthesize a sinusoidally oscillating reference voltage 351having a frequency f_(synth)=f_(ref)=292 Hz corresponding to a puresinusoidal D4 frequency 350. At over 70 kHz, the clock frequency 354 iswell into the ultrasonic range and is therefore not a source of unwantedspectral contamination. Compared the prior art spectrum of square wavegenerated 292 Hz, i.e. a pulsed D4, shown in FIGS. 12A and 12B, theharmonic spectra 353 of the 3^(rd), 5^(th), 7^(th) through 13^(th)multiples of a 292 Hz sinusoid all have zero energy—meaning all spectralcontamination in the audio band has been completely eliminated (seeTable 355).

Aside from being outside the audio range, the magnitude of the noisegenerated by clock frequency 354 is small. A close-up view 352 ofsinusoid 350 shown in FIG. 19C reveals the origin of the noise,incremental steps in voltage 359 present in the generated waveform 358occurring each time the output of the D/A converter changes voltage. Asshown, these transitions occur at the oscillating frequency of the clockused to load the decoder of the DAC. This frequency occurs at afrequency f_(clock)=f_(ref)·(# of DAC steps) where the “# of DAC steps”corresponds to the bit resolution of the D/A converter (rounded to anyconvenient number of steps), although it is also possible to employclock frequencies higher than this clock frequency.

Unless higher clock frequencies are intentionally utilized, thefrequency of the clock and hence the frequency of the noise generated bythe clock will scale with frequency of the sine wave being generated. Assuch, if the sine wave being generated is at a lower frequency, thenoise spectrum of the clock will correspondingly occur at lowerfrequencies, possibly overlapping the audio spectrum. For example, graph360 a shown in FIG. 19D illustrates a portion of a 18.25 Hz sine wave361 comprising a sequence of small voltage changes 362 occurring at theclock frequency of the D/A converter, specifically 4,380 Hz.

On the same time scale, graph 360 b of FIG. 19D illustrates in histogram363 the ΔV_(ref) change in voltage at each of these steps as apercentage of the oscillation's 1.2V peak-to-peak magnitude. Prior to13.7 msec when the output voltage V_(ref) is still increasing, the valueof ΔV_(ref) is positive. At 13.7 msec the change diminishes to near zeroand thereafter the change become negative in polarity. At approximately27.4 msec when the sine wave passes through its average voltage of 0.6V,i.e. a point 364, the magnitude of ΔV_(ref) reaches its largest negativevalue and thereafter begins to diminish in magnitude. This peakmagnitude represents less than 1.3% of the amplitude of the sine waveitself.

The resulting spectra are shown in FIG. 19E, which indicates that themagnitude of the voltage transitions occurring at the clock frequency4,380 Hz, represented by column 367, are small compared to the magnitudeof the oscillating reference voltage at 18.25 Hz, represented by column366. Similarly, the harmonics of these digital transitions are alsonegligibly small in relative magnitude. For example, the magnitude ofthe 3^(rd) harmonic of the clock frequency is represented by the column368. Even though the clock and its 3^(rd) and 5^(th) harmonics are inthe audio spectra, i.e. lower in frequency than 22,000 Hz shown by line175, their small magnitude makes the spectral contamination of thesynthesized oscillating reference insignificant, even at lowfrequencies. If necessary, moreover, the remaining ripple, howeversmall, can be filtered out by the Filter Enable function biased toconnect capacitor 342 to the V_(ref) output by turning on MOSFET 340.

By employing analog synthesis as disclosed herein, a wide range of sinewave excitation patterns in the audio spectrum can be generated to driveLED arrays for phototherapy applications, free from harmoniccontamination. Using the disclosed methods and apparatus in analogsinusoidal synthesis, dynamic control of waveforms in both frequency andin amplitude may be realized including independent control in peak andaverage current control.

As illustrated in FIG. 20, these various combinations are exemplified ingraph 370 a, which shows an Enable signal 371 and reference currentwaveforms 375-379, and graph 370 b, which shows the resulting LEDcurrent waveforms 385-389. These sinusoidal waveforms, summarized in thefollowing table, are not shown to imply a specific therapy or protocolbut simply to illustrate the various current waveform combinationspossible using analog synthesis.

Peak Min Ave Time, Waveform Description En αI_(ref) αI_(ref) αI_(ref)Freq I_(LED) <t₁ Large-signal high-frequency on I_(r1) 0 I_(r1)/2f_(ref0) ΔI_(L1) ± ΔI_(L1) sinusoid 375, 385 with no DC offset t₂-t₁Large-signal low-frequency on I_(r1) 0 I_(r1)/2 f_(ref1) ΔI_(L1) ±ΔI_(L1) sinusoid 376, 386 with no DC offset t₃-t₂ Reduced-signallow-frequency on I_(r2) 0 I_(r2)/2 f_(ref2) ΔI_(L2) ± ΔI_(L2) sinusoid377, 387 with no DC offset t₄-t₃ Small-signal high-frequency on I_(r2)I_(r4) (I_(r2) − I_(r4))/2 f_(ref3) I_(LDC) + ΔI_(L3) ± ΔI_(L3) sinusoid378, 388 with DC offset >t₄ Small-signal low-frequency on I_(r2) I_(r4)(I_(r2) − I_(r4))/2 f_(ref4) I_(LDC) + ΔI_(L3) ± ΔI_(L3) sinusoid 379,389 with DC offset

Graphs 370 a and 370 b are broken into 5 time intervals, with adifferent waveform example in each interval, the intervals before timet₃ representing large signal behavior, where the LED current oscillateswith a peak-to-peak variation that represents a significant fraction ofthe peak available supply current, and the intervals after t₃representing a small variation in current relative to the peak availablesupply current and relative to the average DC current I_(LDC)+ΔI_(L3).Furthermore, the frequencies f_(ref0) and f_(ref3) in the intervalsbefore t₁ and between t₃ and t₄ are shown to be high compared to thefrequencies of the waveforms in the other intervals.

Specifically, in time intervals 0 to t₁ and t₁ to t₂, the magnitude ofreference current waveforms 375 and 376 oscillate between zero and apeak current value of I_(r1) with an average current of I_(r5)=I_(r1)/2shown by dashed line 380 and with respective frequenciesf_(ref0)>f_(ref1). This reference current results in an LED currentΔI_(L1)±ΔI_(L1) having an average LED current ΔI_(L1) illustrated bydashed line 390, a peak current 2ΔI_(L1), and a minimum current of zero.In the subsequent interval from t₂ to t₃, the large signal referencecurrent waveform 377 decreases in peak magnitude compared to theprevious intervals but still remains large signal, with a referencecurrent ranging from zero to I_(r2) with an average valueI_(r6)=I_(r2)/2 illustrated by dashed line 381. Consequentially, LEDcurrent 387 oscillates sinusoidally from zero to a peak current 2ΔI_(L2)around an average current of magnitude ΔI_(L2) represented by dashedline 391. While the frequency f_(ref2) of waveforms 377 and 387 can bechosen to any value, as shown it remains the same as the prior intervalt₁ to t₂, namely f_(ref2)=f_(ref1).

At time t₃ and thereafter the amplitude of the reference currentwaveforms 378 and 379 is reduced dramatically, waveforms 378 and 379ranging between currents I_(r2) and I_(r4) symmetrically around anaverage current of magnitude I_(r3) represented by the dashed line 392and oscillating at frequencies f_(ref3)>f_(ref4) combined with aconstant DC offset I_(r4). The resulting LED currents 388 and 389oscillate sinusoidally at frequencies f_(ref3) and f_(ref4)respectively, and both have an peak-to-trough range of 2ΔI_(L3) and anaverage current represented by dashed line 392, which is equal to a DCoffset I_(LDC) plus one-half the peak-to-trough range 2ΔI_(L3) of thewaveforms 388 and 389, i.e. I_(LDC)+ΔI_(L3). The resulting small signalwaveform therefore is a current oscillating sinusoidally between maximumand minimum values of I_(LDC)+ΔI_(L3)±ΔI_(L3), meaning that the LEDs arecontinuously illuminated but with sinusoidal variation in theirbrightness.

In conclusion, to create time varying currents of regular periodicityfor phototherapy it is preferable to vary the LED current using acontrolled current source or controlled current sink instead of drivingthe LED string with a controlled voltage source, because LED brightnessvaries in proportion to current in a consistent manner. In contrast, LEDvoltage varies in a manner independent of brightness and primarily as aconsequence of variations in LED die fabrication and manufacturing.Maintaining consistency and uniformity of LED brightness using voltagedrive therefore remains problematic, requiring precision trimming ofeach channel of LED drive.

As shown previously, to implement a controlled current sink, aprogrammable voltage is fed into a network of resistors and transistorsto establish a reference current and to mirror this current to one ormultiple channels driving separate LED strings. The value of thereference current may be actively trimmed during manufacture to set theprecise value of current for a given voltage input by trimming a networkof resistors as shown previously in FIG. 16C or by trimming a network oftransistors as shown in FIG. 16D. The transistors may comprise eitherbipolar or MOSFET type.

By varying the voltage used to drive the current mirror ortransconductance amplifier over time in a regular periodic manner, atime dependent or oscillatory LED current may be created. The voltagemay be varied sinusoidally or by any other regular periodic function byoperating the voltage reference in an oscillatory circuit. Alternativelythe voltage can be constantly changed using digital control of avoltage-output type DAC to “synthesize” the desired waveforms.

An alternative means by which to produce a controlled voltage is to feeda time varying programmable voltage into a transconductance amplifier,an amplifier which naturally converts a voltage into a correspondingcurrent, but transconductance amplifiers are larger and more expensiveto implement than using current mirrors.

Still another alternative, at least theoretically, could be to bias eachcurrent sink MOSFET to operate in the constant current regime ofoperation, precisely driving it with the proper gate voltage for eachdesired drain current. To accomplish this goal, the gate drive circuitrywould require calibration at the time of manufacturing. Once calibrated,driving the MOSFET gate with time-varying sequence of voltages willresult in a desired periodic current waveform. Because, however,threshold voltage varies not only with manufacturing but withtemperature, the calibration method to produce controlled and wellmatched currents across multiple channels of LED drive remainsproblematic. As such a current mirror is still vastly superior becausethe two or more mirror transistors vary with manufacturing and overtemperature in the same way so that the current ratio of the transistorsand the resulting LED current remains constant.

Finally, a programmable current-mode DAC can be employed to synthesize aperiodic time varying current, but to drive multiple LED strings, itstill is beneficial to feed the DAC output current into a transistorcurrent mirror not only to buffer the current to a higher value but toconveniently produce multiple channels of well matched LED drive.

Analog Sinusoidal Synthesis of Chords

Referring again to the resonant graph of FIG. 10, it is well documentedthat many if not most physical systems exhibit more than one resonantfrequency. Given the plethora of time constants present in the anatomyand cytological processes of living creatures, it is clear that multiplebioresonant frequencies exist in nature as well. While it is unprovenwhether the simultaneous excitation of multiple bioresonant frequencieshas a beneficial impact on treatment efficacy, prior art systems utilizedigital pulse excitation of the LEDs. As shown in FIG. 11 and FIG. 12,such purely-digital square-wave LED drive methods are incapable ofsimultaneously producing multiple frequencies, except for unwantedharmonics.

In dramatic contrast, it is well known that sine wave frequencies addalgebraically without limitation, as evidenced by the existence ofmultiple note “polyphonic chords” in an acoustic piano. Mathematically,the sum of sine waves can be expressed by the series sum of multiplesine waves of varying magnitude A_(x), frequency ω_(x) and duration (ordecay rate), namelyG(t)=A ₁(t)·sin(ω₁)+A ₂(t)·sin(ω₂)+ . . . +A _(x)(t)·sin(ω_(x))as represented graphically in FIG. 21 where a 292 Hz sine wave 401 and a584 Hz sine wave 402 are combined to produce a two-note chord shown bywaveform 403. LEDs driven by polyphonic excitation will simultaneouslyand concurrently exhibit multiple frequencies, with the ability toeffectively couple energy into comparable bioresonant frequencies.

One means by which to synthesize polyphonic chords in shown in FIG. 22Acomprising an analog mixer circuit 405 summing oscillating referencevoltages V_(refa) and V_(refb) produced by oscillators 236 a and 236 b,respectively, to produce a time-varying voltage resulting in oscillatingreference current αI_(ref) as an input to MOSFET driver 215 a. A largevariety of analog mixer circuits exist including multiple inputamplifiers using adjustable resistor dividers to vary the gain ofindividual inputs. Oscillators 236 a and 236 b, having differentfrequencies of oscillation, may be synchronized to prevent unwantedfrequency drift and aliasing.

Other analog sources may be used to generate a polyphonic referencecurrent comprising one or more chords or even music. For example, inFIG. 22B the analog output of any polyphonic audio source 408 includinga music synthesizer, radio decoder, or audio recording player may beused to generate the reference current αI_(ref), provided that theanalog voltage output of the audio source 408 and series resistance ofthe circuit are adjusted to limit the peak value of αI_(ref) to theinput range acceptable for MOSFET driver 215 a to prevent signaldistortion. Conceptually, the analog voltage output of audio source 408may be scaled in voltage by a voltage divider including resistors 407 aand 407 b followed by audio preamplifier 406 to produce the time varyingcurrent αI_(ref). One way to implement such a circuit is to employ afixed reference current of value I_(ref) and to scale this current to ahigher or lower current with a current amplifier having current gain α,where the gain α is modulated in response to the analog output of analogaudio source 408. The analog audio source 408 may comprise a tapeplayer, a digital audio player, a CD player, or digitally streamedmusic.

Another method, shown in FIG. 22C, to derive an analog audio source isto directly translate a digital source 413 such as digital streamedaudio, digitally encoded data, or a CD audio and to convert the specificdata encoding format into a parallel or serial digital data using formatconversion in an audio codec 412. This stream of 1-bit data or sequenceof 16-bit parallel words is then processed using custom algorithms in adigital signal processor (DSP) 411 and loaded at regular intervals intoa D/A converter 410 to create the desired time varying reference currentαI_(ref). To avoid audio distortion, digital words should be loaded intoD/A converter 410 at a minimum frequency of 44 kHz if the entire audiospectrum is to be preserved.

One common point of confusion is that digital audio sources such as CDplayers or Internet streamed digital audio are often considered digitalbecause the audio information is stored as “bits”, specifically digitalwords describing a sequence of audio volumes commonly referred to as PCMor pulse coded modulation. During reconstruction of the analog audiosignal, however, the digital PCM source is used to drive a D/A converterto produce a time-varying analog signal and as such, signalreconstruction comprises “analog” synthesis in a manner similar to themethod shown in FIG. 22C.

Aside from those similarities, the function of digital audio players isto reproduce an audio signal driving a magnetic coil or piezoelectriccrystal to move air and produce sound, not to produce light. The mass ofa speaker or transducer acts as a natural filter, its inertiaresponsible for removing many unwanted frequencies and spikes. Forexample, combined with a filter capacitor, the inductance of a speakercoil naturally forms a simple low-pass filter. In short, audioreproduction favors low frequencies and has to be driven with highcurrents produced by means of power amplification, in order tofaithfully reproduce high frequency tones. In many cases, such as guitaramplifiers, the amplifier is intentionally driven into distortion aslong as the harmonics sound “good”.

In contrast, photons are massless and not subject to inertial damping orfiltering. The response time of an LED occurs with nanosecond precisionand faithfully reproduces all the harmonics of the driving waveform,even when those harmonics are unwanted or detrimental to the purpose ofphototherapy. As a consequence of these differences, the harmonicspectral content used for driving LEDs in phototherapy is key toachieving bioresonance with specific biophysical process such aselectron conduction, ionic transport, molecular bonding, transientthermal conduction, and steady-state heating of cells, tissue andorgans, regardless of whether analog or digital synthesis is used isgenerating the waveforms for the phototherapy.

For example, when adapting an audio source or music for LED drive, DSP411 may be used to selectively filter certain frequencies and notes froman audio stream while suppressing other tones that may be adverse tophototherapeutic treatment, for example odd harmonics created by cymbalcrashes. Therefore, the data rate at which D/A converter 410 is loadedwith new data should be equal to no less than twice the highestfrequency being reproduced as LED current modulation by MOSFET driver215 a. As a matter of convenience D/A converter 410, DSP converter 411,and audio codec 412 may be synchronized by a common digital clock signal414, often generated by dividing down the oscillations of a crystal(xtal) oscillator. While digital filtering may make music and tonesreproduced on a speaker or headphone sound unlistenable to the humanear, removing unwanted harmonics and spectral content from LED drivewaveforms in phototherapy is important in achieving tissue specificityand high treatment efficacy during phototherapy treatments.

Another inventive method disclosed herein to avoid the complexity andadded costs of analog signal processing, digital filtering, or audiomixing to produce chords of tones, is to combine an analog synthesizedwaveform with a second digital pulse frequency achieved by digitally“strobing” an analog oscillating waveform. Returning to the circuit ofFIG. 17B, such a method employs the single frequency oscillator 236 tofeed the reference current input of MOSFET driver 215 while strobing theMOSFET driver on and off using digital synthesizer 203 a. Two possiblemethods exist, namely

-   -   setting the digital strobing frequency f_(clock) to be higher        than the frequency of the oscillating reference current f_(ref),        i.e. f_(clock)>f_(ref)    -   setting the frequency of the oscillating reference current        f_(ref) to be higher than digital strobing frequency f_(clock),        i.e. f_(ref)>f_(clock)        The waveforms produced using these two methods have differing        spectral characteristics, and therefore the methods cannot be        used interchangeably to perform dual-frequency LED drive.

FIG. 23A illustrates the case wherein the frequency f_(clock) of theclock signal is higher than the frequency f_(ref) of the sinusoidallyoscillating reference current, i.e. the first of the methods describedabove. As shown in graph 420 a, a 292 Hz oscillating sinusoidalreference current 421 (D4) with a period T_(ref)=3.42 msec and anaverage value 422, clearly has a longer period and lower frequency thanthe digital pulses of an Enable signal 423 having a clock periodT_(clock). For the purposes of this illustration, the specific frequencyf_(clock) of the digital pulses of Enable signal 423 may be any valueprovided that f_(clock) is at least double the sine wave frequencyf_(ref). During operation, MOSFET driver 215 outputs zero volts, i.e.ground, whenever Enable 423 is at a logic zero and the analog value ofoscillating reference current 421 whenever Enable signal 423 is a logicone or “high” state. The resulting waveform is equivalent to multiplyingthe analog sine wave by the digital multiplier of “1” or “0” for eachmoment of time, essentially “chopping” a sine wave into pieces.

The LED current waveform shown in graph 420 b comprises small pulses ofcurrent of varying height, where the collection of pulses forms anenvelope 425 a, 425 b, 425 c, or 425 d (individually and collectively as425) having the same frequency and phase as oscillating referencecurrent 421. The difference of these envelopes is a variation only inamplitude depending on the ratio of t_(on) to T_(clock) of Enable signal423. The duty factor of the Enable signal 423, i.e., t_(on)/T_(clock),acts as PWM brightness control, controlling the average current of thesinusoidal envelope 425 and hence LED brightness by pulse widthmodulation, without changing the frequency or phase of the sinusoidalreference current 421.

Because the higher of the two frequencies in the polyphonic chord iscreated “digitally”, this frequency component will exhibit thepreviously described harmonics of a square wave, contributing tounwanted spectral contamination. This point is illustrated in FIG. 23B,where the 292 Hz reference current 421 occurs at a frequency f_(ref)shown by line 431, and combines with the 4,672 Hz digitally pulsedEnable signal 423 that occurs at a frequency f_(clock) shown by line432. Because the Enable signal 423 is a square wave, it producesharmonics 434 including a 3^(rd) harmonic at 14,016 Hz in the audiospectrum and the remainder of its harmonics in the ultrasonic spectrum,i.e. beyond the frequency illustrated by line 175. So using this method,a chord of 292 Hz (D4) and 4,672 Hz (D8) can be generated without theneed for a mixer or two analog oscillators, with the only disadvantagethat an unwanted 3^(rd) harmonic is still present in the audio range.The resulting spectra are summarized in table 435 including otheroctaves of D for reference.

If the digital pulse rate is increased to D9 or any other note higherthan approximately 7 kHz, no harmonic will be manifest in the audiospectrum. This example is shown in FIG. 23C, where the 292 Hz referencecurrent shown by line 431 is combined with a 9,344 Hz digitally pulsedEnable signal 441 at a frequency f_(clock) (D9) shown by line 440. Theresulting spectra are summarized in table 445 including other octaves ofD for reference.

Note that, as shown in FIG. 23D, if the clock frequency f_(clock) shownby line 450 can be pushed into the ultrasonic spectrum, then as shown intable 451 no harmonics of concern exist. Because this method eliminatesthe second note of the chord it is therefore not a method for polyphonicsynthesis and confers no advantage over leaving the Enable signal oncontinuously. As an alternative, running the clock at 18,688 Hz, i.e. atD10, as shown by solid line 452 in FIG. 23E, eliminates all audioharmonics but still offers a second frequency as an octave of f_(ref).

In summary, for polyphonic synthesis of two tones wheref_(clock)>f_(ref), there is no restriction of the value of f_(ref) butthe digital pulse generated frequency f_(clock) must be chosen to avoidsignificant spectral contamination in the audio range.

FIG. 24 illustrates the case where the Enable signal is digitallystrobed at a frequency f_(clock) that is lower than the frequencyf_(ref) of the sinusoidally oscillating reference current, i.e. wheref_(clock)<f_(ref).

As shown in graph 460 a, a fixed-frequency constantly oscillatingreference current 462 with period T_(ref) and average value 464oscillates with longer period and lower frequency than digital pulses ofEnable 461 having a clock period T_(clock). Each clock period Tclock issubdivided into two intervals—t_(off) when Enable 461 is at a logic zeroor biased in an “off” condition, and t_(on) when Enable 461 is biased ata logic one or “high” state. During operation, MOSFET driver 215 outputszero volts, i.e. ground, whenever Enable 461 is at a logic zero.Conversely, whenever Enable 461 is a logic one or “high” state, MOSFETdriver 215 outputs the time varying analog values of oscillatingreference current 462.

During this t_(on) interval, the output of the MOSFET driver 215 a doesnot result in a single, constant LED current but whatever portion of thesinusoidal oscillation in voltage and current is occurring at that time.The resulting waveform is equivalent to multiplying the analog sine waveby the digital multiplier of “1” or “0” for each moment of time,essentially “chopping” the sine wave into short intervals or “snippets”of oscillation. The LED current waveform shown in graph 460 b comprisesthe same intervals of duration t_(on), where the LED current 466completes one or several oscillating cycles before it is is shut off, asshown by line 467, for a duration t_(off) and thereafter repeating theentire cycle.

In the event that reference current waveform 463 includes a DC offsetwith an average value 465, as shown in graph 460 a, then the resultingLED current waveforms 468, shown in graph 460 b, exhibit identical ACoscillatory behavior, except that the magnitude of the oscillation isreduced, resulting in oscillatory perturbations in brightness of an LEDstring that repeatedly conducts for a duration t_(on) and thentemporarily is interrupted for a duration t_(off) before resuming itsconduction and small signal oscillations. Note that the absence orpresence of a DC offset in the oscillatory reference current has noimpact on the harmonic spectra of the two-note chord.

The resulting spectra for a chord of D8 and D9 using this disclosedmethod is shown in FIG. 25A, where a sinusoidal reference current at afrequency f_(ref) of 9,344 Hz (D9), shown by solid line 472, combineswith an Enable signal digitally pulsed at a frequency f_(clock) of 4,672Hz (D8), shown by solid line 423. Because the Enable signal is digitallypulsed, it produces harmonics 434, including a 3^(rd) harmonic at 14,016Hz in the audio spectrum and higher-frequency harmonics in theultrasonic spectrum, i.e. beyond the frequency illustrated by line 175.So, using this method, a chord of D8 and D9 can be generated without theneed for a mixer or two analog oscillators, with the only disadvantagethat an unwanted 3^(rd) harmonic is still present in the audio range.The resulting spectra are summarized in table 473, including otheroctaves of D for reference.

While this method works fine for high frequency chords, its operation isproblematic when generating low frequencies because the digital clock,the origin of harmonic noise and spectral contamination, occurs at thelower frequency of the two-note polyphonic chord. This problem isillustrated in FIG. 25B, which shows the result of mixing a 584 Hz (D5)reference current (line 476) with a 292 Hz (D4) digitally pulsed Enablesignal (line 161). Because of the 292 Hz square wave Enable signal,spectral contamination of harmonics 164 occur throughout the audiospectrum, as described in Table 477, identical to those shown in FIG.12. Such a method is therefore not useful for generating low frequencypolyphonic chords for LED drive in phototherapy applications.

For generating high frequency polyphonic chords, the method can beimplemented at low cost as shown in FIG. 26 because the oscillator 236used to create the sine wave can also be used to drive a simple divideby 2, 4 or 8 counter 482 to simply generate the digital clock pulsesneeded as the Enable signal input to MOSFET driver 215 a. Because theoscillating reference 236 exhibits sinusoidal transitions too slow forclearing triggering counter 482, an intervening Schmidt trigger orcomparator 481 with hysteresis and high input impedance is insertedbetween the oscillator 236 and the counter 482. Each factor of 2 infrequency division implemented by counter 482 represents an octave inmusical notes, e.g. D8 divided by 2 is D7, D8 divided by 4 is D6 and soon.

Pulse Width Modulated Digital LED Control

In addition to analog sinusoidal synthesis described above, anotherinventive means disclosed herein to synthesize sinusoidal waveforms withcontrolled harmonic content for driving LEDs in a phototherapy system isthrough the use of digital synthesis. While analog synthesis involvessinusoidally varying the reference or bias current to the LED currentcontrol circuit, digital synthesis involves pulsing the LED currenton-and-off in constantly varying durations to synthesize a sine wave (orchords of multiple frequencies of sine waves). Pulse modulationtechniques include both fixed-frequency “pulse width modulation”,commonly referred to by the acronym PWM, and variable-frequency “pulsefrequency modulation”, referred to by the acronym PFM. While both PWMand PFM modulation techniques may be employed to control average currentor voltage in electronics circuits, such as voltage regulators, thevariable clock rate of PFM complicates waveform synthesis. Moreover, PFMcan give rise to unwanted radio frequency noise and electromagneticinterference (EMI) which varies in frequency and is therefore difficultto filter or shield. EMI is especially problematic in medical devicesbecause government agencies such as the FDA and the FCC strictlyprohibit EMI that could dangerously interfere with other life-criticalmedical devices in a hospital or clinic. As a consequence, the digitalsynthesis section of this application mainly focuses on PWM controltechniques with the understanding that, alternatively, a sequence of PFMcontrolled pulses may be used in waveform synthesis of so desired.

Returning to the waveform examples shown in FIG. 15, while the pulseddigital waveforms 243 through 258 do not specifically illustrate digitalsinusoidal synthesis, the ability to change the average LED current froma level shown by dashed line 272 to a higher level 273 simply byincreasing the LED current pulse width 267 to a longer pulse width 268.Since the frequency of both of pulses 267 and 268 is equal to 1/T₁, thisrepresents the principle of “pulse width modulation”, also known asfixed-frequency PWM, one means by which to perform sinusoidal synthesisdigitally. The alternative method of digital synthesis, “pulsedfrequency modulation” or also a “PFM”, is exemplified by comparingpulses 268 and 269 at times t₈ and t₉ used to increase the average LEDcurrent from a level shown by dashed line 273 to 274 by varying the LEDon-time and frequency, i. e., since T₂ is greater than T₁, the frequencyof pulse 268 (1/T₁) is greater than the frequency of pulse 269 (1/T₂).Variable frequency PFM methods may comprise fixed-on time or fixed-offtime modulation schemes. Variable frequency PFM methods are oftenavoided because of concerns of time-varying signals contributing todynamically changing electromagnetic interference resulting in noisethat is difficult to filter.

Unlike analog circuits whose performance and circuit stability aresensitive to electrical loading of their outputs, in digital synthesis,the enable signal produced by the digital synthesizer circuitry has alarge digital “fan-out,” meaning that one digital synthesizer can beused to control many channels and MOSFET drivers. An example of a largefan-out is illustrated in FIG. 27C where digital synthesizer 203 has asingle output and is used to drive the Enable input of numerous MOSFETdrivers from 215 a through 215 n, where n is a variable representing thenumber of MOSFET drivers. In this example, where digital synthesizer 203has a single output, all the channels of LED drivers will exhibit thesame digital waveform and synthesize the same sinusoids synchronously.This centralized approach allows one digital synthesizer to connect toall the MOSFET drivers using a shared conductive signal path, whether awire, conductive printed circuit board (PCB) trace, or a data bus.

FIGS. 27A, 27B, and 27C illustrate and contrast various combinations ofdigital synthesizers and independent channels of LED drive. In FIG. 27A,each MOSFET driver 215 a through 215 n is controlled by its owncorresponding digital synthesizer 203 a through 203 n (collectively asdigital synthesizer 203), where the subscript “n” is a mathematicalvariable representing the number of MOSFET drivers and digitalsynthesizers. These various digital synthesizers shown may occupy one,several, or completely independent integrated circuits representingeither a centralized, clustered, or fully distributed system. Becauseeach LED channel and associated MOSFET driver are controlled by theirown dedicated digital synthesizer, this implementation offers completeflexibility in synthesizing sinusoids of channel-unique frequency,magnitude, and duration should it be desired. As such, it is importantthat the channels be synchronized to a common clock reference, or noisemay result from channel-to-channel interactions and aliasing. In thisindependent and autonomous approach, each of digital synthesizers 203a-203 n must connect to its corresponding one of MOSFET drivers 215a-215 n with a dedicated wire or conductive PCB trace.

Another method which minimizes duplication of circuitry and minimizes ICreal estate without sacrificing flexibility is a centralized method ofcontrol shown in FIG. 27B comprising a single digital synthesizer 203having multiple independently controlled outputs. In this approach, thecentralized digital synthesizer 203 must uniquely address every MOSFETdriver with a separate and distinct wire or conductor. If discrete wiresor conductive PCB traces are employed, the digital synthesizer must belocated near, i.e. in the physical vicinity of, the MOSFET drivers orotherwise a large number of conductors of extended length will berequired. Alternatively a data bus may be employed to distribute thedata for all channels, but then each channel requires a decoder circuitto uniquely identify its particular control signal from the others.

One implementation of the digital synthesizer 203 a of FIG. 27A isschematically represented in FIG. 28A, comprising a digital counter 503,a latch 506, and a digital buffer string comprising inverters 507 a and507 b, with the output of digital synthesizer 203 a controlled by clocksignals 501 and parallel data bus 502 generated by microcontroller μC500. Inverters 507 a and 507 b are shown to illustrate that the outputof latch 506 comprising minimum size logic transistors must be bufferedto drive the input capacitance of one or more MOSFET drivers 215 a, aswell as to compensate for any parasitic resistance and capacitancepresent in the conductive interconnect between digital synthesizer 203 aand an electrical load, represented by current sink circuit 201 a. Assuch, the current drive capability and the corresponding gate width ofthe MOSFETs used in inverter 507 b should be sized accordingly to drivethe Enable line at the requisite speed.

While the illustration shows a single inverter 507 a electricallyinserted between the un-buffered output of latch 506 and the input tohigh current inverter 507 b, in practice many intermediate inverters ofsequentially increasing gate widths (not shown) may be used to scaleeach inverter's output current with the capacitive loading of the nextinverter. So long as the total number of inverters in the series ofinverters (including the first inverter 507 a and the last inverter 507b) is an even number, e.g. 2, 4, 6, . . . then the output of digitalcounter 503 and latch 506 should remain in digital phase with the outputof digital synthesizer 203 a. The result of employing the describedsequential buffer string is a significantly larger fan out and abilityto consistently drive a wide range of data lines while contributing anegligible change in signal propagation delay. Throughout thisdisclosure, this same technique may be used anytime a high-speed gateneeds to drive a long line, high capacitance, or heavy load at a highspeed, and therefore it will not be described again.

In operation, μC 500 writes data from its pattern EPROM onto paralleloutput lines 502. μC 500 also generates clock signals on lines 501,comprising a Sync pulse and clock signal θ. In operation, a Sync pulsesets the output of latch 506 to logic “1” which, buffered by inverters507 a and 507 b enables MOSFET driver 215 a into an on state, drivingthe gate of MOSFET 216 a to produce a programmed current I_(LED) andilluminating LED string 205 a to a fixed brightness. Concurrently, theSync pulse causes digital counter 503 to load the data present onparallel data bus 502 into the counter's register 504, shown by exampleas an 8-bit word. Pulses of clock signal θ cause digital counter 503 tocount down linearly, decrementing the remaining count by one with eachpulse. When the count reaches zero, digital counter 503 generates apulse on output line 505, resetting the output of latch 506 to “0” anddisabling MOSFET driver 215 a.

The timing diagram of FIG. 28B illustrates digital synthesizer operationof digital counter 503 in graph 510 a and operation of latch 506 ingraph 510 b. As shown, digital counter 503 loads data 512 upon loadinstruction 511 triggered by the Sync pulse on one of clock signal lines501. Repeated pulses of the clock signal θ subsequently decrement thecounter register 504 once for each interval T_(θ), eventually countingdown to zero count at time 513. During this time, the output of thedigital synthesizer 203 a outputs a logic “1” state as shown by waveform516. When the value of the count in digital resister 503 reaches zero,the output is reset (line 517) and the LED string switches off at time513. Until the next load pulse (line 511), the count in digital counter503 remains at zero (line 514) or alternatively is ignored even if itcontinues to count.

As illustrated, digital counter 503 is binary and may comprise a ripplecounter or a synchronous counter. Alternatively the counter 503 may berealized by software within μC 500, eliminating the need for hardwarecounters and latches, but still performing similar functions. Inconclusion, the PWM counter function within digital synthesizer 203 amay be implemented discretely, or using a dedicated timer functionwithin μC 500, or implemented in software within μC 500. When softwaretimers are employed, however, care must be maintained to insure thatinterrupts do not suspend or delay regular counter operation, or anincorrect frequency may be synthesized.

The resulting LED current waveforms of the disclosed LED drive systemcomprise pulses of controlled widths and varying duration repeated at afixed clock rate. By varying the on time t_(on) while maintaining afixed clock period T_(sync), the average current in an LED string can becontrolled digitally. Such a method can be referred to a fixed-frequencypulse width modulation or PWM control. Examples of fixed-frequency PWMgeneration of pulses of varying on-time are illustrated in FIG. 28C. Inphototherapy applications, PWM average current control can be used fordynamic brightness adjustment of digitally pulsed LED currents as shownin FIG. 8B and described in previously cited U.S. Pat. No. 9,877,361.Alternatively, such PWM methods disclosed herein can be used for digitalsynthesis of sinusoidal waveforms, driving LED strings in an inventivemanner free from spectral contamination in the audio spectrum.

Unlike in analog synthesis described in the previous section, where theaverage LED current is changed by altering the conducted LED currentusing a sinusoidal reference voltage, in digital sinusoidal synthesis asequence of pulse widths varying in a prescribed manner are employed torecreate the sinusoidal waveform at a frequency far below the clock rateused to generate the pulses themselves. As shown in FIG. 28C, pulse 520comprises an on-time t_(on50) which is half that of the clock periodT_(sync), specifically with a digital value of “1” for the portion 520of the waveform and with a digital value of “0” for the remainingportion 521 of the T_(sync) period. As such the on-timet_(on)=50%·T_(sync), the off-time t_(off)=1−t_(on)=50%·T_(sync), and inthis particular case t_(on)=t_(off).

The average current during any PWM pulse is determined by its dutyfactor, defined as D≡t_(on)/T_(sync). Accordingly, in this example theduty factor is given by D=t_(on50)/T_(sync)=50%, where dashed line 522graphically illustrates the duty factor, visually representing theaverage value of the waveform. Starting with waveform 523, the top rowof waveforms shown in FIG. 28C illustrates pulses 524 with duty factorsgreater than 50%, specifically duty factors of 61%, 71%, 79%, 82% and99%. In the 99% waveform, the dashed line 526 representing the averagevalue and the off-time shown by line segment 525 are not drawn to scalein order to better illustrate the variables. Similarly, starting withwaveform 527, the bottom row of waveforms shown FIG. 28C illustratepulses with duty factor less than 50%, specifically duty factors of 39%,29%, 21%, 18% and 1%. In the 1% waveform, the dashed line 529representing the average value and the on-time shown by pulse 528 arenot drawn to scale in order to better illustrate the variables. Eachexample in the top row is located above its complementary waveform inthe bottom row, i.e. the mirror image condition around the 50%condition. For example, waveform 524 with an on-time t_(ton61) and a 61%duty factor has a duty factor 11% above the 50% center value, whilewaveform 527 with an on-time t_(ton39) and a 39% duty factor has a dutyfactor 11% below the 50% center value.

By stringing together, i.e. sequencing, a series of pulses of varyingduty factors and fixed period in a specific manner, any mathematicalfunction, including sinusoidal waveforms, can be generated from PWMmodulated digital pulses. For example, in FIG. 29A, a series of digitalpulses 590 of varying width, e.g. t_(on50), t_(on82), t_(on21), etc.occurring at a fixed period T_(sync) results in a time-varying averagevalue synthesizing a pure sine wave 592. During this digital synthesis,the value of analog reference current 591 remains constant and does notcontribute to generation of the sine wave. In this method, sine wave 592can be synthesized to have any frequency and period independent of theclock frequency 1/T_(sync), provided that the clock frequency 1/T_(sync)is higher than the highest frequency 1/T_(synth) being synthesized.

Provided that the clock frequency f_(sync)=1/T_(sync) is chosen to benear or greater 22 kHz, neither the digital clock frequency nor itsharmonics are present in the audio spectrum, and the resulting digitalsynthesis produces no spectral contamination that could adversely impactphototherapy efficacy. For example, a 28,032 Hz clock can be employed tosynthesize a 1,168 Hz (D6) sine wave with 24 independent T_(sync) timeintervals. Such an approach is equivalent to breaking a 360° sine waveinto 24 pieces of 15° and 35.7 μsec each as illustrated in graph 600 ofthe digital synthesizer's normalized magnitude versus time shown in FIG.29B. Plotting the average value of sine wave 601 against elapsed timeexpressed in fixed 15° angle increments 602 results in a spectrumcomprising the frequency f_(synth)=1/T_(synth) of the generated sinusoid602 along with the clock frequency f_(sync)=1/T_(sync) used to generateit. In pulse width modulation, the magnitude of each pulse determined bythe PWM duty factor has the same average amplitude as that of a D/Aconverter with the same resolution.

Unlike a D/A converter, however, in PWM control the actual analog valueis not present in the amplitude of a waveform but in its durationdetermined by the time average value of the current or voltage. Thisduration is illustrated by waveforms 604 a through 604 d having PWM dutyfactors of 50%, 100%, 75% and 25% corresponding to arc angles of 0°,90°, 150° and 330° respectively. The average value 602 of any 15° timeincrement comprises a portion of time when the output is at the fullscale of 100% and a the remainder of the period where the output is at0%. The average value shown as sinusoid 600 is in between, varying inproportion to the duty factor of each time slice.

As a practical matter, in sinusoidal synthesis using digital circuitry,negative voltages are problematic because they require dual power supplyvoltages, e.g. ±0.6V, where the signal must range from voltagesabove-ground to those “below-ground”. Negative or below ground voltagesare uncommon in integrated circuits, difficult to integrate because theyrequire special electrical isolation techniques, and almost unheard ofin digital circuitry. To realize a sinusoid using only positive supplyvoltages, the average value of the sine wave must occur above ground.For example, if sine wave 601 is realized using 1.2V logic, then for asinusoid having a peak-to-peak voltage range of 1.2V, i.e. ±0.6V, theaverage voltage of the sine wave occurs at 0.6V. In digital synthesisthis center voltage occurs at D=50%, equivalent to the zero state of asine wave occurring at 0°, 180°, and at 360°.

A direct comparison between analog synthesis and fixed-frequency PWMdigital synthesis of a sinusoid is shown in FIG. 29C, where the verticalaxis represents the amplitude of the synthesized sine wave in a giveninterval while the horizontal axis represents time within the interval.In analog synthesis using a D/A converter (DAC), the amplitude of thesignal shown in graph 620 a controlled by the DAC output remains at aconstant voltage for the entire period T_(sync). In any given interval,the normalized DAC output has a value V_(on)/1.2V ranging from 0% to100% and may vary in the next time increment by a change in magnitude622. These magnitude changes generally comprise linear steps of ±ΔV,±2ΔV, etc. according to any desired resolution comprising 256 levels foran 8-bit DAC, 4096 levels for a 12-bit DAC and 65,536 steps for a 16-bitDAC. Since the instantaneous voltage of the waveform is set by the DACand not by a PWM counter, then the highest required clock frequency toimplement analog synthesis is 1/T_(sync) with the period T_(sync)adjusted in accordance with the highest frequency to be reproduced withfidelity.

In contrast, using PWM digital synthesis, in a plot of voltage versustime shown in graph 600 b, at the beginning of each time interval thevoltage jumps from 0% up to 100% with no intermediate values except fortransitions, and remains at this voltage for some fractional t_(on) time625 of the T_(sync) period 627. The on-time t_(on) is dynamicallyadjusted in linear increments of time±Δt, ±2Δt, etc. set by a 8-bit,12-bit, or 16-bit counter having a resolution of 256, 4096 or 65,536steps respectively according to the desired resolution unless otherwiselimited by available clock frequencies. Because the average value of thesinusoid is set by a clock counting time and further subdividing thepulse shown in graph 600 b, then a higher clock rate is needed tosynthesize a sine wave than is required using a D/A converter. So, whileanalog synthesis achieves its resolution with steps in voltage, PWMdigital synthesis achieves its resolution by steps of time. As such, themaximum frequency of the clock required for PWM digital synthesis is1/T_(θ) where this frequency is the sync clock frequency 1/T_(sync)times the desired resolution desired. In PWM synthesis, each timeinterval, e.g. 604 a, includes a portion of the time current is flowingin the LED and a portion of time where the drive current is zero.Provided the clock frequency f_(sync) is sufficiently high to be beyondthe audio spectrum, then the cells in living tissue cannot respond tothe presence of this high frequency, especially since it represents asmall signal change in the average current from one interval to thenext. In essence the cells provide natural filtering. Another filteringeffect occurs because of capacitance in the LEDs and the MOSFET drivecircuit which unavoidably softens the driving current waveform edges andfilters high frequency noise, particularly harmonics beyond the audiospectrum, Lastly additional capacitance can be added to the LED drivechannels if required.

Sinusoidal reconstruction with good fidelity, i.e. sinusoidal synthesiswith minimal harmonics from distortion of the waveform from itsmathematically ideal shape, requires a sufficient number of intervals ofthe highest sinusoidal frequency being reproduced f_(synth)(max). Foranalog synthesis this clock frequency f_(sync) is given by the relationf _(sync)=1/T _(sync)=(#intervals)·f _(synth)(max)where the variable “#intervals” is the number of time intervals per 360°for the highest frequency waveform being synthesized and f_(synth)(max)is the highest frequency waveform being synthesized. One means by whichthe #intervals can be chosen is by the desired width of each timeinterval in degrees using the following relation: #intervals=360°/(arcangle of each time interval). For example if each arc angle is 36° then#intervals=10, if each arc angle is 20° then #intervals=18, if each arcangle is 15° then #intervals=24, if each arc angle is 6° then#intervals=60, and so on. This hyperbolic relationship, that smallerangles require more time intervals to describe one full 360° cycle of asine wave, means in PWM synthesis higher resolution, requires a fasterclock.

To summarize the comparison, digital PWM synthesis requires a higherfrequency clock f_(θ) than analog synthesis because each time intervalT_(sync) must be further subdivided into smaller snippets of time ofduration T_(θ), meaning for digital PWM synthesis the same bitresolution requires a higher clock frequency than analog synthesis. Therequired frequency f_(θ) of this faster clock, the one used for countingthe increments of the on-time and setting the duty factor, is given bythe relation

$\begin{matrix}{f_{\theta} = {1/T_{\theta}}} \\{= {( {{bit}\mspace{14mu}{resolution}} ) \cdot f_{sync}}} \\{= {( {{bit}\mspace{14mu}{resolution}} )/T_{sync}}} \\{= {( {{bit}\mspace{14mu}{resolution}} ) \cdot ( {\#\;{intervals}} ) \cdot {f_{sync}( \max )}}}\end{matrix}$essentially describing how many thin rectangles of fixed time intervalsare used to reconstruct one cycle of the highest frequency to besynthesized. This faster PWM clock signal f_(θ) may be generated from aneven higher fixed frequency oscillator f_(osc), preferably temperaturecompensated to minimize drift, using either a constant or dynamicallyadjustable frequency ratio. The process of dividing the synthesizedsinusoidal waveform into small rectangles of fixed duration and ofheight equal to the magnitude of the function is analogous to themathematical procedure called “integration” in calculus. In integralcalculus, as the time increments “dt” become infinitely thin, thesynthesized waveform is reproduced precisely and the area under thecurve, the energy and harmonic content of the phototherapy excitation,is precisely controlled. Also it should be noted that the value ofT_(sync) is identical for both analog and digital synthesis. Forexample, using 18 intervals of 20° each to synthesize a 1,168 Hz (D6)sinusoid, the Sync clock used to load D/A converters in analog synthesisor to load the digital counter in digital PWM synthesis has a frequencyf_(sync) of 21,024 Hz, a frequency sufficiently high that it and all itsharmonics occur at the extreme upper range of the audio frequency rangeand beyond.

Graph 640 a in FIG. 29D illustrates a plot of the clock frequencyrequired in the system as a function of the maximum frequency sine waveto be synthesized, shown ranging from D4 to D8. The y-axis representsthe highest frequency clock which in the case of analog synthesisrepresented by line 641 is the sync pulse used to load the D/A converterat a frequency of f_(sync) and in the case of digital PWM synthesis isthe digital counter clock having a frequency f_(θ). Using digitalsynthesis of the same 1,168 Hz waveform, the digital clock rate of thePWM digital counter for 8-bit, and 10-bit resolution shown by lines 642and 643 respectively requires corresponding clock frequencies f_(θ) ofapproximately 5.38 MHz and 21,529 MHz. For 12-bit resolution, thedigital counter clock is 4,096 times that of f_(sync) or over 86 MHz,too high to be shown on the graph.

Graph 640 b also shown in FIG. 29D illustrates the linear impact ofincreasing the number of time intervals used to synthesize 360° of thehighest frequency sine wave being generated, where the number ofintervals varies from 8 to 30. As shown by line 645 the clock raterequired to synthesize a 2,336 Hz (D7) sine wave remains below 5 MHz foremploying a 6-bit counter offering 64 magnitudes for a 1.2V sine wave,i.e. where each step represents 18.8 mV or 1.6% increments of thesignal. Line 646 illustrates an 8-bit counter offering 256 steps and aprecision of 4.69 mV or 0.4% step increments can be achieved over thefull range without exceeding 20 MHz.

Considering that practical commercial microcontrollers typically operateat clock frequencies between 10 and 25 MHz, line 647 illustrates that a10-bit PWM counter can only be used with a small number of intervals, 8or less, while remaining below 25 MHz. Using fewer than 12 intervals per360° results in distortion in the synthesized sinusoid not compensatedfor by higher bit precision, meaning the benefit of more preciselysetting the average voltage in a given time interval by using 12-bit PWMcounters or larger, is not worth sacrificing the number of timeintervals used to construct the sinusoid. For high fidelity synthesis ofa sinusoid free from unwanted audio spectrum harmonics, the number oftime intervals for practical considerations ranges from 12time-intervals each 30° wide, to 24 intervals of 15°. The followingtables details the clock frequency required to synthesize a 4,672 Hz(D8) sinusoid using a various sized PWM counters.

bit accuracy 8 × 45° 12 × 30° 15 × 24° 18 × 20° 24 × 15°  6-bits  2.4MHz  3.6 MHz  4.5 MHz  5.4 MHz  7.2 MHz  8-bits  9.6 MHz 14.4 MHz 19.9MHz 21.5 MHz 28.7 MHz 10-bits 38.3 MHz 57.4 MHz 71.8 MHz 86.1 MHz  115MHzOf the above conditions, the shaded boxes are not viable either becausethe clock frequency exceeds 25 MHz or because the number of timeintervals are too few. This analysis suggests that the optimum conditionis a 21.5 MHz PWM clock driving a 8-bit PWM counter to synthesize a4,672 Hz (D8) sinusoid from 18 time intervals, each 20° in width. Thecorresponding PWM clock has a period T_(θ)=1/f_(θ)=1/(21.529 MHz)=46.5nsec and sync period of T_(sync)=256/f_(θ)=11.9 μsec with acorresponding frequency f_(sync)=83.9 kHz.

While discrete oscillator solutions can be utilized, in many cases theaccuracy and cost is unwarranted, especially considering that many suchsolutions were developed for radio communications. On the other hand, 25MHz oscillators are relatively easy to manufacture discretely or inconjunction with common microcontrollers because this oscillatingfrequency is commonly used in Ethernet communications. One timing sourceand clock generator circuit 660 made in accordance with this inventionis illustrated in FIG. 30, comprising oscillator 661, digital counters662 and 664 and trim register 693 to create clock signals 501 used todrive the digital synthesizer 203 a shown in FIG. 28A.

Oscillator 661 may be realized using a crystal oscillator, an R-Crelaxation oscillator, a ring oscillator, or a silicon MEMs oscillator.A crystal oscillator, comprising a crystal shard of quartz mechanicallytuned to resonate a specific frequency is advantageous for itstemperature independence, but it is unfortunately relatively fragilecompared to semiconductors. An R-C relaxation oscillator employs aresistor-capacitor network to charge the capacitor at a set rate,discharging the capacitor rapidly after reaching a comparator or Schmidttrigger threshold, and repeating the process interminably. In manycases, the circuit elements to implement timing source 660 are fullyintegrated into μC 500 (shown in FIG. 28A) and are entirelyuser-programmable in firmware or software.

Clock precision is achieved by trimming the resistor in an R-Coscillator and/or using materials that are relatively temperatureindependent. Another alternative is the to create a time source using alarge number of inverters connected head-to-tail, i.e. output to input,to form a loop or ring. When powered, the signal propagates around theinverter ring at a frequency in accordance to the inverters' propagationdelays. An odd number of inverters are required to insure theoscillations continue. The newest solution available today is the use ofsilicon micromachine devices or MEMs, used to create a small vibratingspring or diving board (cantilever) monitored electrically by capacitivecoupling or peizo-resistive variation and tuned to resonate according toits specific mass.

Regardless of the technique employed, the oscillator 661 produces a 25MHz oscillating signal which is then adjusted to any lower desiredfrequency, e.g. 21.5 MHz, by digital counter 662. If oscillator 661 istrimmed during manufacturing then counter 662 can be preset to a fixedvalue by software. If however, the frequency of oscillator 661 varieswith manufacturing, functional trimming using trim register 663 isnormally performed during manufacturing. In functional trimming,measurement of frequency f_(θ) is made repeatedly while the count beingloaded into counter 662 by the digital value stored in trim register 663is adjusted until the desired frequency is achieved and the frequencysource calibrated.

This PWM clock frequency is supplied to the digital synthesizer and alsoto the input of programmable counter 664, converting the PWM clockfrequency f_(θ) into the Sync pulse having a frequency f_(sync) that is,as shown, 256 times lower than f_(θ). The divide by factor for counter664 should match the desired resolution of the PWM output, e.g. 8-bits,10-bits etc. In this manner the PWM digital counter 664 will countpulses corresponding to the frequency f_(θ) and the Sync pulse occurring256 pulses later will reset the LED driver and restart the count.

As applied to LED drive in phototherapy, the effective resolution ofsinusoidal generation using the disclosed invention can be estimated bymultiplying the number of time intervals used in constructing thesinusoid times the number of PWM duty factors possible, i.e. the bitresolution of the PWM counter. Multiplying 18 time increments,approximately equivalent to 4-bit precision, times 256 possible valuesof D generated from an 8-bit counter means for sinusoids up to 5,425 Hz,the total resolution is approximately equivalent to 12-bits or 4096combinations. Unless the clock frequency is increased in proportion tof_(synth)(max), using PWM methods to synthesize sinusoids above thisfrequency means the aggregate resolution must be reduced hyperbolically,i.e. where f_(osc)/f_(synth)(max) sacrificing fidelity either bylowering the bit-resolution or the number of time intervals. Thistradeoff between the maximum frequency synthesized and its aggregateresolution is illustrated in the following table:

description D8 bandwidth D9 D10 ultrasonic waveform f_(synth)(max) 4,672Hz 5,425 Hz 9,344 Hz 18,688 Hz 22,000 Hz oscillator f_(osc) 25 MHz 25MHz 25 MHz 25 MHz 25 MHz f_(osc)/f_(synth)(max) ratio 5,351 4,608 2,6761,338 1136 waveform resolution 4,608 4,608 2,676 1,338 1136 equivalent12-bit 12-bit 11-bit 10-bit 10-bit resolutionThe table illustrates that for synthesizing sinusoids up toapproximately 5.4 kHz, the overall resolution of the digital synthesizeris 4608 combinations, greater than 12-bit resolution. Above thisfrequency, referred herein as the synthesizer's “bandwidth”, the digitalsynthesizer's resolution declines in proportion to the sinusoid'sfrequency, declining to 11-bit precision at 9,344 Hz (D9) and maintainsat least 10-bit resolution all the way to the upper edge of the audiospectrum. The bandwidth limitation and its impact is illustratedgraphically in FIG. 31 wherein curve 671 shows the aggregate synthesizerresolution versus the maximum synthesized frequency f_(synth) (max) inboth the number of possible combinations and in their bit equivalence.As shown, the accuracy of digital synthesizer 203 a remains constant ata value exceeding 12-bits until the frequency of 5.425 kHz, the digitalsynthesizer's bandwidth, is reached (line 673), above which theresolution declines proportionately with f_(synth)(max). At the edge ofthe ultrasonic spectrum (line 175), the digital synthesizer 203 a stillmaintains an overall resolution of 10-bits. If the number of timeintervals used to synthesize the highest frequency sine wave ismaintained at #intervals=18, then the drop in aggregate resolution 671must be accompanied by a decrease in PWM counter resolution as shown byline 672. Even operating above synthesizer 203 a's bandwidth, up to theedge of the ultrasonic spectrum 175, the PWM counter resolution stillexceeds 6-bits.

Clearly, above synthesizer 203 a's bandwidth, as the resolution declinesthe fidelity of the synthesized sine wave suffers. While for audiophileslistening to music, subtle distortion and phase artifacts of the digitalaudio reproduction process may be noticeable to the trained ear, in anLED drive for phototherapy the resulting distortion is essentiallyinsignificant, carrying little energy and occurring at harmonicfrequencies outside the audio spectrum. No adverse impact is expected inthis frequency range.

As described previously, at a frequency slightly above 7 kHz, even thelowest harmonics of a square wave are outside the audio spectrum and notexpected to affect photobiomodulation and phototherapy efficacy. So atfrequencies above the threshold frequency shown by line 673 in FIG. 31the disclosed invention may continue PWM synthesis with reducedfidelity, switch to pulsed digital operation, or switch to analogsynthesis described previously. The resulting harmonic spectra, shown inFIG. 32A, illustrate that using PWM digital synthesis of a sinusoidresults in only the synthesized frequency represented by line 675 in theaudio range. The sync frequency f_(sync) used to load the data streaminto the PWM digital counter, represented by line 676, occurs at afrequency far into the ultrasonic spectrum beyond the upper limit of theaudio spectrum (line 175). The clock pulses used to control the PWMon-time (line 678) and the clock pulses used to generate it (line 677)occur in the MHz range and are not present in the LED drive excitationwaveforms whatsoever.

When the same approach is employed to synthesize a lower frequency sinewave, e.g. f_(synth)=292 Hz (D4) shown by line 681 in FIG. 32B, apotentially serious noise problem results. If the synthesized frequencyof 292 Hz is generated using the minimum required Sync clock frequency(line 682), the resulting clock frequency f_(sync) occurs at 7,078 Hz inthe middle of the audio spectrum and with relatively high energycontent. Moreover as described by table 679 in FIG. 32B, the thirdharmonic of the Sync clock (line 683) also falls at a frequency belowthe lower limit of the ultrasonic spectrum (line 175), in the upper partof the audio spectrum. So while using the minimum possible clockfrequency is beneficial in synthesizing high frequency waveforms, it isnot advantageous in generating lower frequency sinusoids.

As illustrated in FIG. 32C, requiring the upper limitation in PWM clockfrequency f_(θ) not to exceed the preferred oscillator frequency 25 MHz,and the lower limitation in the Sync pulse frequency f_(sync) not tofall within the audio spectrum puts practical constraints on the rangeof frequencies f_(synth) that can be synthesized using the fixed clockratio described previously, namelyf _(θ)=(bit resolution)·f _(sync)=(bit resolution)·(#intervals)·f_(synth)(max)For the PWM clock frequency for synthesized sine waves formed oftwenty-four 15° time intervals or eighteen 20° time intervals to remainat or below 25 MHz, shown by horizontal line 680, the maximum frequencysinusoid f_(synth)(max) is limited to 4,069 Hz and 5,425 Hzrespectively, as shown by points 682 a and 682 b and consistent withFIG. 31. According to the above relation at the other extreme,synthesizing any sine wave having a frequency f_(synth) below 917 Hzwith 15° time intervals or below 1,222 Hz with 20° time intervals meansthat the Sync clock pulse frequency f_(sync) will be sufficiently lowthat it falls below the frequency represented by line 175 and into theaudio band, specifically shown as points 684 a and 684 b, creating thepotential for unwanted spectral contamination affecting phototherapyefficacy. The resulting range bounded by the audio spectrum's limitationof the Sync clock f_(sync) on the low-end and the practical limit of theoscillator's 25 MHz frequency on the PWM clock frequency f_(θ) on theupper end (shown by shaded region 685 for the 20° synthesis example).Assuming the oscillator frequency and audio boundaries are fixed,operating outside of the allowed range means resolution must besacrificed for synthesizing high sinusoidal frequencies and at the otherextreme, a higher than minimum, i.e. an “over-sampled” Sync clockfrequency must be maintained when synthesizing low frequency sinusoids.

In conclusion, when the required clock frequency for PWM digitalsynthesis is impractically high, the options available using thedisclosed invention include

-   -   limit the maximum frequency of the synthesized sine wave    -   compromise the harmonic fidelity of the synthesized waveform by        limiting PWM bit resolution, i.e. reducing the resolution of the        duty factor    -   compromise the harmonic fidelity of the synthesized waveform by        employing larger time intervals, thereby reducing the number of        time intervals per T_(synth)    -   switch from digital synthesis to analog synthesis above a        certain clock frequency, using a D/A converter as described        previously including to vary the magnitude of the LED current in        accordance with analog, digital and PCM sources    -   Combinations of the above methods        Conversely, when the frequency of the sine wave being        synthesized is too low, the minimum Sync clock frequency must be        maintained above a set frequency limit and cannot scale in        proportion to the synthesized frequency. Using the inventive        methods disclosed herein a sinusoid of controlled and        dynamically adjustable frequencies for LED phototherapy can        therefore be generated using digital synthesis free from        spectral contamination of unwanted harmonics in the audio        spectrum.

Digital Sinusoidal Synthesis

Given the aforementioned description of the apparatus and methods ofpulsed width modulation control of LED current, frequency, andbrightness, any sinusoid, series of sinusoids, or chords of multiplesinusoids may be dynamically synthesized.

Referring again to the apparatus of FIG. 28A, in sinusoidal synthesis, aparticular control sequence, i.e. a specific series of PWM counts, issequentially loaded from any digital controller such as μC 500 intoregister 504 of digital synthesizer 203 a. The digital synthesis ofsinusoids in accordance with the methods described herein controls theharmonic content and brightness of one or more LED strings used inphototherapy. While microcontroller μC 500 is shown as the source ofthese instructions, any programmable logic or logic array, customdigital circuitry or custom integrated circuit may also be used togenerate the control sequence.

Whether by hardware, software or some combination thereof, execution ofthe digital synthesis involves a sequence of steps such as those shownin FIG. 33. Starting with the step “Select Pattern” (step 700), the LEDwavelengths, channels, and driving algorithms are chosen. In “LoadConditions” (step 701), these settings including f_(sync), f_(θ),t_(on), T_(sync), T_(synth), and the various synthesis patterns areloaded into the appropriate registers within μC 500 and in associatedhardware, counters, buffers, etc. If a single-frequency sinusoidf_(synth1) is to be synthesized, the sequence of digital codes requiredis recalled from non-volatile memory files and then saved in a dataregister or stack. These codes represent the counts loaded sequentiallyinto the PWM counter each time a T_(sync) pulse occurs. If a chord ofmultiple sinusoids f_(synth1)+f_(synth1)+ . . . +f_(synthx) is to besynthesized, a different sequence of digital codes is recalled fromnonvolatile memory file comprising and loaded into a data register orstack. Data registers may comprise static or dynamic memory, i.e. SRAMor DRAM, but since they are modified, i.e. “written” frequently andrapidly during synthesis, the data registers operate at a frequency toohigh for non-volatile memory such as EPROM, E²PROM or flash, used tostore the phototherapy patterns and algorithms.

After the conditions are loaded into registers or stacks for quickaccess, in “Load Tsync Counter” (step 702 a) the register 705 containingdata that represents the first time interval T_(sync) is loaded into theT_(sync) counter 664, shown in FIG. 30. In tandem, in “Load PWM Counter”(step 702 b), the data in register 706, representing the on-time of thepulse within the time interval T_(sync), is loaded into PWM counter 503shown in FIG. 28A. In the step entitled “Set Latch, Enable LED, CommenceCounting” (step 702 c) the output of PWM latch 506 is set “high”enabling MOSFET driver 215 a and illuminating LED string 205 a.Concurrently, T_(sync) counter 664 and PWM counter 503 commence countingpulses from the f_(θ) clock. In the step entitled “Decrement PWM Counterto Zero” (step 702 d), PWM counter 503 counts down to zero while theT_(sync) counter continues unabated. When the PWM counter 503 reacheszero, the output of PWM latch 506 is reset “low” disabling MOSFET driver215 a and turning off LED string 205 a as described by the step entitled“Reset Latch, Disable LED, Continue T_(sync) Count” (step 702 c). As thename describes, the T_(sync) counter continues to count through the stepentitled “Decrement T_(sync) Counter to Zero” until the Tsync countreaches zero.

Once T_(sync) counter 664 reaches zero, a program decision (step 703) ismade in accordance with the algorithm prescribed by files originallyloaded during the “Select Pattern” step 700. If the pattern has beencompleted in the “Pattern Complete” case (arrow 704 a), the sequence isfinished and a new pattern must be selected to continue. Otherwise, inthe case “Pattern Not Complete” (arrow 704 b) a new set of countscomprising the data in register 705, representing the new time intervalT_(sync) 705, and the data in register 706, representing the on-time ofthe pulse within the time interval T_(sync), are respectively loadedinto T_(sync) counter 664 and PWM counter 503, and steps 702 a through702 f are repeated. The process continues until decision 703 determinesthe pattern is complete, whereby program execution terminates anddigital synthesis of a sequence of sinusoids or sinusoidal chords iscomplete.

In software implementations, the size of counters 702 a and 702 b areadjustable, able to synthesize a single cycle of a sinusoid or multiplecycles. The duty factor of a given pulse may be calculated as the ratioof the on-time determined by the count stored in register 706 and theT_(sync) time interval stored in register 705. While in fixed frequencyPWM synthesis, the T_(sync) time interval in register 705 remainsconstant and the on-time in register 706 is adjusted to control the dutyfactor, the T_(sync) period can be adjusted to synthesize any givensinusoid of an arbitrary frequency f_(synth). The algorithm shown inFIG. 33 accommodates changing the value of T_(sync) in accordance withthe frequency of the sinusoid being synthesized and to maintain adesired resolution. For example, f_(sync) can be decreased in proportionto the maximum frequency f_(synth)(max) of the sinusoid beingsynthesized. Alternatively, a higher value of f_(sync) than required maybe employed.

For example, except for the aforementioned of audio frequency noiseissue, a 292 Hz (D4) sinusoid may be synthesized using an 8-bit PWMcounter and either 24 or 18 time intervals. In graph 730 of FIG. 34A,sinusoid 731 a is synthesized using 24 evenly-spaced intervals eachcorresponding to 15° of arc and having a duration of 140.7 μsec. Eachinterval has an average value shown by steps 731 b determined by an8-bit PWM counter having 256 durations summarized in table 732. Bysuccessively loading the PWM counter with the binary equivalent of thedecimal number in the “PWM count” column or the hexadecimal number inthe “hex” column of table 733, the sinusoidal waveform 731 a willresult. In operation, at the first time point representing 0°, the PWMcounter is loaded with hex number 80 for 50%, the sin of 50°. Because ofa quantization error in the counter, i.e. 128/255, the nearest dutyfactor is 50.2%, the synthesizer exhibiting a slight discrepancy fromits ideal average output. After 140 μsec, one T_(sync) time interval,the PWM counter is loaded with a new value A0 hex (160 decimal) changingthe duty factor to 62.7%.

The process continues sequentially driving the average magnitude highertill at 0.86 ms the PWM counter is loaded with FF hex reaching a dutyfactor of 100%. Thereafter the PWM duty factor declines reaching aminimum value at 2.57 ms of 0 corresponding to the sine of 270°. Theprocess then repeats to synthesize additional cycles of sinusoids. Themajor negative aspect of this sinusoidal synthesis is the noisegenerated by f_(sync)=7,008 Hz shown in table 732. While it does notcomprise an entire spectrum of audio frequency harmonics present inpresent day digital pulsed systems intentionally operating in the audioband, it still represents audio spectral contamination.

In graph 730 of FIG. 34B, sinusoid 736 a is synthesized using 18evenly-spaced intervals each corresponding to 20° of arc and having aduration of 190.3 μsec. Each interval has an average value shown bysteps 736 b determined by an 8-bit PWM counter having 256 durationssummarized in table 737. By successively loading the PWM counter withthe binary equivalent of the decimal number in the “PWM count” column orthe hexadecimal number in the “hex” column of table 738, the sinusoidalwaveform 736 a will result. The advantage of dividing a sine wave into20° intervals of time over that of 15° intervals is the lower resolutionallows a higher frequency sinusoid to be synthesized with a clockfrequency f_(θ). The disadvantage of employing 20° intervals is that thenearest points to the maximum and minimum values on the sinusoid at 90°and 270° occur at 80°, 100°, 260° and 280° causing some flattening ofthe synthesized sine wave, slight distortion appearing as if thewaveform was “clipped”. Another negative aspect of this sinusoidalsynthesis is the noise generated by f_(sync)=5,256 Hz shown in table737. While it does not comprise an entire spectrum of audio frequencyharmonics present in present day digital pulsed systems intentionallyoperating in the audio band, it still represents audio spectralcontamination.

A time graph of PWM pulses 739 used to synthesize sinusoid 736 a and itssequence of average value steps 736 b is shown in greater detail in FIG.34C. For clarity the average value of each step 736 b is listed as apercentage for each interval along with the corresponding decimalequivalent of the binary count loaded into the 8-bit PWM counter.

FIG. 34D illustrates synthesis of a single cycle of 1,168 Hz (D6)sinusoid 741 a with PWM average value shown by steps 741 b comprising 18time intervals of 20°. In this case, the PWM clock frequency f_(θ) andthe sync interval T_(sync) are adjusted from f_(θ)=1.346 Mhz to 5.198MHz and from T_(sync)=190.3 μs to 49.3 μs, commensurate with thedecrease in the period of the synthesized sinusoid from 3.42 ms to 0.86ms as summarized in table 742. The PWM counter sequence used tosynthesize sinusoid 741 a is described in table 743 both in hexadecimalform and its decimal equivalent. Since the Sync frequency isf_(sync)=20,304 Hz, no audio spectrum noise is generated.

FIG. 34E illustrates the same data for synthesizing a 4,672 Hz (D8)sinusoid 746 a shown in graph comprising steps 746 b formed inaccordance with PWM count sequence shown in table 748 and clock periodsshown in table 747. Comparing these conditions with the synthesis oflower frequency sinusoids illustrates that the minimum frequency clockrate requirements for the PWM clock f_(θ) change with synthesisaccuracy, i.e. the number of time intervals used to synthesize thesinusoid (#intervals), and with the frequency of the sinusoid beingsynthesized f_(synth).

Frequency f_(synth) (Note) 292 Hz (D4) 1,168 Hz (D6) 4,672 Hz (D8)Period T_(synth) 3.42 ms 0.86 ms 0.21 ms #intervals (degrees) 18 of 20°24 of 15° 18 of 20° 24 of 15° 18 of 20° 24 of 15° Sync dock period 190.3μs 140.7 μs 47.6 μs 35.7 μs 11.9 μs 8.9 μs T_(sync) Sync clock freq.f_(sync) 5,256 Hz 7,008 Hz 21,024 Hz 28,032 Hz 84,096 Hz 112,128 Hz PWMclock freq. fe 1.35 MHz 1.79 MHz 5.38 MHz 7.18 MHz 21.53 Mz 28.70 MHz

As the above table reveals, the PWM clock frequency f_(θ) increases inproportional to the frequency being synthesized with synthesis at 15°increments carrying a 33% overhead in added clock rate compared to 20°resolution. This added accuracy only becomes limiting when synthesizingthe 4,672 Hz (D8) frequency or higher, because 28.7 MHz exceeds thecommon clock frequency 25 MHz used in microcontrollers and for Ethernet.The table also clarifies that synthesis of a 292 Hz sine wave using theminimum frequency f_(sync) results in noise in the audio spectrum, atapproximately 5 kHz and 7 kHz. This problem can be avoided usingoversampling, discussed below.

While the aforementioned waveforms comprised sinusoids with peak-to-peakamplitudes representing 100% of the digital scale, the magnitude of thesynthesized sine wave can be reduced simply by changing the sequentialPWM code, as shown in table 753 in FIG. 35A. In the digital synthesizedwaveform 751 shown in graph 750, the average value of the function is+25% and varies with an amplitude 754 of ±25%, ranging in total from 0%to 50%, i.e. with a sinusoidal output of 25%±25%. Without changing theoperating conditions in table 752 from that of a full-scale sinusoidspecified previously in table 732, the magnitude and the mean value ofthe digitally synthesized sinusoid can be controlled simply by adjustingthe PWM code sequence labeled “Hex” in table 753 to lower magnitudenumbers.

Although this reduced magnitude sine wave shown in FIG. 35A extendeddown to 0% at its minimum, as shown in FIG. 35B, even with a reducedmagnitude sinusoid of ±25% shown by line 764, the entire curve can beshifted up by a DC offset 765, in this example by +25%, to produceresulting offset sinusoid 761 with a DC bias offset. In phototherapythis waveform modulates the LED brightness while maintaining someillumination at all time. The shift is the average value and the smallermagnitude of the oscillation is accomplished entirely by minimizing thevariation in the sequential PWM code described in table 763.

As FIG. 35C reveals, modification of the PWM code as shown in table 773can be used to further limit the AC swing to a small signal level, e.g.±10% variation. This AC component 774 can be considered small signalwhen compared to the DC component 765 of the waveform 771, comprising a+60% offset 765 in the entire sinusoid. The resulting spectrum is shownin FIG. 35D illustrating a sinusoid of limited amplitude (line 781) atfrequency of 1,168 Hz (D6) (line 780). As graphically represented, thesinusoid of limited amplitude (line 781) sits atop a DC offset (line782). By definition, direct current or DC has a frequency of zero Hertz.The Sync clock has a frequency (line 783) of 28 kHz, well outside theaudio spectrum.

Digital Sinusoidal Synthesis of Chords

An LED phototherapy drive system made in accordance with this inventionis also capable of digitally synthesizing chords of multiple frequenciesfor driving LED strings. When more than one frequency pattern ispresent, e.g. a higher-frequency sine wave of period T_(synth1) and alower-frequency sine wave of period T_(synth2), the duration of thepattern is chosen to synthesize at least one cycle of the lowerfrequency. This means the overall time of the pattern has a duration ofat least T_(synth2) and over the same interval more than one 360° cycleof the higher frequency sinusoid will necessarily occur. Assuming forsimplicity's sake that the ratio of the sinusoids is an integer, i.e.where T_(synth2)=βT_(synth1), then more than β cycles of the higherfrequency sinusoid will occur is the same time that only one cycle ofthe lower frequency sinusoid occurs. For example, a single cycle of a1,168 Hz (D6) sine wave requires 0.856 ms to complete 360° while a 4,672Hz (D8) sine waves requires only 0.214 ms. The ratio of their sinusoidalperiods is therefore β=4, meaning four complete cycles of the 4,672 Hz(D8) sine wave is completed in the same time interval that only onecycle of the 1,168 Hz sinusoid is completed.

An example of this higher frequency component is shown in FIG. 36 wherean individual cycle of a 4,572 Hz sinusoid having a periodT_(synth1)=0.214 ms is repeated four cycles having a total period forthe pattern synthesized of βT_(synth1)=4T_(synth)=4·0.214 ms=0.856 ms.The resulting curve 801 shown in graph 800 comprises the same pattern ofsynthesized duty factor and digital PWM codes described in table 803 afor the duration from 0 to 0.214 ms and then repeats in columns 803 b,803 c, and 803 d for the corresponding time intervals from 0.214 ms to0.428 ms, from 0.428 ms to 0.642 ms, and from 0.642 ms to 0.856 ms. Alltold, synthesis of four cycles of a 4,672 Hz sinusoid requires4·0.214=0.856 ms to complete, comprising 4·18=72 time intervals.

In order to accurately add two or more waveforms together to form achord in digital synthesis disclosed herein, each function must have adefined value at the same time points, even if the value must beinterpolated from other time points. For example to add the values of a1,168 Hz sine wave together with that of four cycles of a 4,672 Hz sinewave 801, both sine waves must have a corresponding value at each timeincrement of 0.214 ms. So while synthesis of one 360° cycle ofhigher-frequency sine wave 801 will comprise only 18 time intervals, thelower frequency sine wave will comprise 72 time intervals, many morethan required for its high-fidelity synthesis. Synthesis of a waveformwith more time intervals than is practically needed for high fidelityreproduction is herein referred to as “oversampling”.

An example of an oversampled sinusoid is illustrated in FIG. 37Acomprising a 1,168 Hz sinusoid 811 generated from the PWM average value812 of 72 distinct time intervals, 4× the number needed to faithfullysynthesize sinusoid 811 with high fidelity. The benefit of oversamplingincludes

-   -   reducing output ripple    -   simplifying filtering of high frequency clock signals    -   preventing the Sync clock frequency from falling into the audio        spectrum when synthesizing low frequency sinusoids    -   increasing the resolution to include common time points where        the amplitude of two or more sinusoids of differing frequencies        may be added to digitally synthesize a chord of frequencies.

For example, in pattern tables 815 a, 815 b, and 815 c shown in FIG. 37Bdefining the PWM counts used to synthesize sinusoid 811, only the shadedrows are needed to synthesize the waveform with fidelity. The rest ofthe PWM counts represent oversampled data. Since only one-in-four PWMcounts are needed to accurately produce the desired sine wave, the drivedata is 4×, i.e. four-times, oversampled.

In this case, such a waveform can be directly added together withsinusoid 801 of FIG. 36 to produce a new waveform comprising a chord oftwo sine waves. The process of adding waveforms to produce a newwaveform comprising a chord of the two component frequencies is showngraphically in FIG. 38 where graph 820 a illustrates the two componentfrequencies of the chord, namely one cycle of 1,168 Hz (D6) sinusoid 811and four-cycles of 4,672 Hz (D8) sinusoid 801, each equal in amplitudehaving a peak-to-peak amplitude of 100% and an average duty factor of50%. While 4-cycle sinusoid 801 has a period T_(synth1)=0.21 ms shown byline 821, lower frequency sinusoid 811 has a period T_(synth2)=0.86shown by line 822, four times longer than T_(synth1). Because the twocurves are integral multiples of one another, oversampling facilitateseasy addition of the PWM counts at each time interval in order tosynthesize the chord of the two notes.

The resulting composite frequency representing a chord of the componentfrequencies is shown by waveform 823 in graph 820 b in FIG. 38. Thesinusoidal nature of the waveform and its constituent frequencies arenot easily identified from the time waveform shown in graph 820 b. Inthe frequency spectrum shown in FIG. 39, however, it can readily be seenthat the synthesized frequencies represented by lines 828 and 827 equalto the 6^(th) and 8^(th) octaves of D are of equal amplitude and theonly synthesized frequency below the upper limit of the audio spectrum(line 175). The sync clock occurs at a frequency 18 times that of thehighest frequency, i.e. 18·4,672 Hz=84,096 Hz (line 829) well into theultrasonic spectrum.

As more notes are added to the chord or if the constituent frequencieshave different amplitudes, the waveform becomes even more complexvisually. An example of mixing sinusoids of differing frequency andamplitude is illustrated in graph 830 a of FIG. 40 where a 1,168 Hz (D6)sinusoid 811 having a peak-to-peak amplitude of ±50% around a 50%average value is mixed, i.e. algebraically added, to 4 cycles of a 4,672Hz sinusoid 831 having an attenuated AC magnitude 852 of ±7.5%, withsinusoid 831 sitting atop a DC offset 833 of +17.5%, meaning sinusoid831 ranges from a low value of 17.5% to an upper value of 32.5%. Inphototherapy, a DC offset can be interpreted as a minimum current andcorresponding brightness which an LED will never drop below. Theresulting waveform 834 from the summation of the two sinusoids into achord is shown in graph 830 b of FIG. 40. Despite the fact that waveform834 and waveform 823 of FIG. 38 both comprise identical frequencycomponents and harmonic spectra, specifically the notes of D6 and D8,the time waveforms appear entirely different.

A process by which the pattern tables used for sinusoidal synthesis,e.g. tables 815 a through 815 c, are created involves an algorithm shownin FIG. 41 or some modification thereof. In this method, starting withthe number of time intervals, e.g. #intervals=18, then the arc degreecolumn of data is calculated using a fixed angle namely Φ=360/18=20°.The column arc degrees Φ, combined with the frequency of the synthesizedwaveform f_(synth), e.g. f_(synth)=4,672 Hz, results in a calculatedtime interval T_(sync)=1/T_(synth)=(20°/360°)/4,672 Hz=0.012 ms. Giventhe foregoing, if the number of cycles β=1, the total period βT_(synth)is then βT_(synth) 1·(18·0.012 ms)=0.214 ms. The result is time intervaltable 843 comprising a column of angles versus corresponding timepoints. If two cycles are desired, i.e. number of cycles β=2, then theheight of time interval table 843 is doubled where the time columnextends from 0 ms to 0.428 ms in increments of 0.012 ms and thecorresponding arc angle ranges from 0° to 720° in increments of 20°.

The time interval table 843 of time versus arc angle Φ is next processedline-by-line by normalized mathematical function 840, in the example bysinusoid function [A·(sin(Φ)+1)+B]≤100%. As indicated, the function isnormalized, i.e., represented as a percentage from 0% to 100%. Arepresents the amplitude and B the offset of the sine wave. Theamplitude A is calculated from the vertical midpoint between thepeak-to-peak values of the sine wave; the offset B is calculated fromthe minima of the sine wave. Thus, 0>A≤0.5 and 0≤B<1, and when A=0.5,B=0. The result is analog sine table 844 comprising columns of time withcorresponding arc angle Φ and the output of normalized mathematicalfunction 840, the exact normalized value of the sine function at eacharc angle, provided the function does not exceed 100%.

For example, in the un-scaled sine wave with no DC offset shown in FIG.34D, the multipliers A=0.5 and B=0 so that the output of normalizedmathematical function 840 is [0.5·(sin(Φ)+1)+0] having values rangingfrom 0% to 100% with an average value of 50%. In the case of anattenuated sine wave with a scaled amplitude A=0.25 and no DC offset B=0as shown in graph 750 of FIG. 35A, the output of normalized mathematicalfunction 840 is [0.25·sin(Φ)+1)+0] and ranges from 0% to 50% with anaverage value of 25%. In the case of an attenuated sine wave with a DCoffset as shown in graph 750 of FIG. 35B, A=0.25 and B=0.25, the outputof normalized mathematical function 840 is [0.25·(sin(Φ)+1)+0.25] havingvalues ranging from 25% to 75% with an average value of 50%. In theexample shown in graph 770 of FIG. 35C illustrating a highly attenuatedsine wave with a large DC offset, A=0.10 and B=0.60, whereby the outputof normalized mathematical function 840 is given by[0.10·(sin(Φ)+1)+0.60] with values ranging from 60% to 80% and anaverage value of 70%.

In the event that the calculated value of normalized mathematicalfunction 840 exceeds 100%, e.g. [(A·sin(Φ)+1)+B]>100%, then the outputof mathematical function 840 is pinned at a 100%, the maximum value ofthe function. In such cases the top portion of the waveform will“clipped” at a maximum value of 100%, and the resulting waveformdistortion will likely produce unwarranted harmonics and spectralcontamination. For stimulating healing in phototherapy where spectralcontrol and the prevention of unwanted harmonics is important, thepreferred LED excitation pattern is a distortion free sinusoidalwaveform with even harmonics. In other cases such as photodynamictherapy, i.e. using photons to excite or chemically activate a chemicalcompound or pharmaceutical, or in efforts to target cellular destructionof bacteria or viruses, other waveforms may also be beneficial. Themathematical operation performed by normalized mathematical function 840may therefore represent any time varying and preferably cyclical,function and is not limited to sinusoids. Regardless of the function, itis convenient to scale the analog output of this operation to “exactvalues” ranging between 0% to 100%, i.e. normalized data. Whilenormalization is not actually required, limiting the data range byscaling and normalization to a range of 0% to 100% makes subsequent dataprocessing of the analog table 844 more convenient in avoiding signalsgreater than the input range of any subsequent mathematical operations.

The term “exact values” for the purposes of this disclosure meansgreater accuracy than the LSB, i.e. the least significant bit of thedigitization process in subsequent steps of the pattern generatingprocess. The resulting output includes an analog duty factor rangingfrom 0% to 100%. In the event that the sinusoid has an attenuatedamplitude A<50%, e.g. A=25%, results in an output that is limited to arange of duty factors less than full scale.

Referring again to FIG. 41, analog sine table 844 is then inputted intoan analog-to-digital converter 841, wherein each percentage value of thefunction (A·sin(Φ)+1)+B is converted into an equivalent digital dutyfactor to later be used in a PWM counter to generate sinusoids. Theconversion process is chosen to match the bit resolution of the intendedPWM counter. For example, in digitizing the analog output of normalizedmathematical function 840 using 8-bit conversion for use in an 8-bitcounter, the duty factor is a digitized value or count ranging from 0 to255 in decimal format shown in digitized sine table 845. The data mayalso be represented by a hexadecimal equivalent of this count rangingfrom 00 to FF, but in actual use, the PWM counter operates digitallyusing base-2 Boolean logic. The process of digitization naturally roundsthe exact analog value to its nearest digital equivalent value, the PWMcount with an analog value closest to the original analog value input toanalog-to-digital converter 841.

The decimal equivalent of the analog value stored in analog sine table844 is then loaded into PWM counter emulator 842 to generate thequantized output “synthesized duty factor” a key component of patterntable 846 used to synthesize sinusoids in real time. The synth dutyfactor column in pattern table 846 represents the analog synthesizedvalue closest to the original exact value in analog sine table 844, thesmall difference being the digitization error resulting by theconversion process of analog-to-digital converter 841. This error can bereviewed when creating pattern table 846 to determine if the agreementwith the original is acceptable. If not, a higher bit resolution may beused with the caveat that the maximum frequency of the synthesizedsinusoid may be reduced by employing higher resolution data conversion.While the decimal equivalent of the duty factor is used to drive the PWMcounter controlling LED drive, the analog value in pattern table 846 isuseful to drive display graphics.

While the algorithmic process to generate a pattern file shown in FIG.41 can be performed in real time “on the fly” or in advance, it isbeneficial to perform the process in advance for commonly usedfrequencies and to store the collection of pattern files in a “patternlibrary” for convenient access during normal machine operation inphototherapy treatments.

In the same manner, chords of two or more sinusoids can be generated inreal time or made in advance and stored in the pattern library as shownin the algorithm of FIG. 42A. In this process the time interval table isgenerated from the input conditions for both sinusoid A having frequencyf_(synthA) and sinusoid B having frequency f_(synthB). The number oftime intervals and hence the gradation of arc angle Φ must be chosen tomeet the minimum acceptable number of intervals on the higher frequencysinusoid. To add the amplitudes of different frequency sinusoids the twosine waves should have the same time scale. As a consequence, the lowerfrequency sine wave will be oversampled such as the one shown in FIG.37A, having a greater number of time intervals and a finer gradation ofarc angles Φ than is required for synthesis with high fidelity. Eachtime-interval table is then converted into exact values of magnitudeG(Φ) using normalized mathematical functions 850 a and 850 b and outputin their corresponding analog sine tables (not shown) wherebyG _(A)(Φ)=[A·(sin(Φ)+1)+B]_(A)G _(B)(Φ)=[A·(sin(Φ)+1)+B]_(B)corresponding to two sine waves of differing frequencies.

These amplitude values are then scaled by scalar multipliers 851 a and851 b C_(A) and C_(B). After scaling, the magnitudes are addedarithmetically together with any DC offset C_(DC) using arithmetic logicunit (ALU) 851 or equivalent programs to facilitate a weighted-sumaddition of the component analog waveform data outputted from thenormalized mathematical function generators 850 a and 850 b. Theweighted average of these waveforms in ALU 852 is given byWeighted Average={C _(A) ·G _(A)(Φ)+C _(B) ·G _(B)(Φ)+C _(DC)}/(C _(A)+C _(B) +C _(DC))In the event that C_(A)=C_(B)=1 and C_(DC)=0, then the WeightedAverage={G_(A)(Φ)+G_(B)(Φ)}/2 and the output is the average of the twovalues. In the case of a weighted average, e.g. where C_(A)=2 andC_(B)=1, sinusoid A contributes twice as much to the chord as sinusoid Bdoes, in which caseWeighted Average={2G _(A)(Φ)+G _(B)(Φ)}/3If a DC offset comprising a quarter of the maximum amplitude is addedthe signal, the above equation becomesWeighted Average={2G _(A)(Φ)+G _(B)(Φ)+1}/4After mixing, the output of ALU 852 is then digitized usinganalog-to-digital converter 853, resulting in the signal magnituderepresented by a digital code used to control the on-time of a PWMcounter. To complete the chord pattern table 855, the digital code isconverted by PWM counter emulator 854 back into an analog valuerepresenting the duty factor. The only error introduced by this processis the single digitization error that occurs from rounding the weightedaverage output of ALU 852.

Because numerical errors are introduced only once, i.e. when generatingthe chord pattern file, the algorithm of FIG. 42A offers superioraccuracy. This accuracy is especially beneficial when synthesizingcomplex pattern files for inclusion in a pattern library and used laterfor subsequent playback. One disadvantage of the algorithm is complexityintroduced by numerical weighted averaging of multiple analog values andrequiring subsequent digitization, making it less amenable to real timesynthesis of chords than purely digital signal reconstruction methods.

An alternative approach using purely digital reconstruction to createchords, shown in FIG. 42B, utilizes the algorithm described in FIG. 41to generate individual sinusoidal pattern files using normalizedmathematical function A 860 a and analog-to-digital conversion 861 a tocreate sinusoid A pattern table 862 a and similarly using normalizedmathematical function B 860 b and analog-to-digital conversion 861 b tocreate sinusoid B pattern file 862 b. These individual pattern tablescan be saved in digital form in the pattern library and used later forgenerating chords.

As shown in FIG. 42B, to generate a chord, the individual sinusoidpattern tables 862 a and 862 b are scaled, i.e. multiplied digitally byE_(A) digital multiplier 860 a and E_(B) digital multiplier 860 brespectively. These scaled files are then added digitally to the digitalE_(DC) DC offset 863 c and added using Boolean algebra in ALU 864, whoseoutput is converted into a synthesis chord pattern by PWM counteremulator 854. Alternatively, the data can be fed directly into a PWMcounter to provide real time control of LEDs.

One complexity of digital chord synthesis is creating files wherein themathematical function of the composite waveform is continuous inamplitude and in slope, i.e. in its 1^(st) derivative, from the end ofone pattern and the beginning of the next pattern. This goal is mosteasily addressed by sinusoids having composite frequencies that areintegral multiples of one another, i.e. where β is an integer, asillustrated in the examples of FIG. 43. In all the examples, the lowerfrequency sinusoid 870 is combined with higher frequency sinusoids 872,873, 874, 875, 876 and 878 representing higher frequencies that areintegral β multiples of the frequency of sinusoid 870, specificallywhere β equals 2, 3, 4, 5, 6, and 8, respectively.

Because the frequencies of the sinusoids 872, 873, 874, 875, 876 and 878are integral multiples of the frequency of the sinusoid 870, each of thesinusoids begins and ends at the same value, namely D=50.2%. The reasonthe duty factor is 50.2% rather than 50% is an artifact of thedigitization process. Even through the PWM counter has 256 levelsincluding 0 volts for a zero code, the number of maximum intervals is255 steps, i.e. that 255 represents 100%. So code 128 is not exactlyhalf of 255 steps, but instead is 128/255=50.2%

As such, a chord comprising any mix of these two component frequencieswill have the same amplitude at the beginning and end of the synthesizedpattern and when repeated sequentially will form a piecewise continuouswaveform in amplitude and in its 1^(st) derivative function. Inaccordance with the prior discussions of even harmonics and theirimportance in phototherapy efficacy, even multiple sinusoids 872, 874,876 and 878 are preferred. The sinusoids 872, 874, and 878, specificallybeing multiples of two of the frequency of sinusoid 870, representoctaves of the fundamental.

In the event that component frequencies of a chord have a ratio that isnon-integral, using a pattern comprising a single cycle of the lowerfundamental frequency will not achieve a continuous function acrossrepeated patterns. Any discontinuity across repeated patterns causes asudden jump in LED current and results in unwanted harmonics, harmonicspresent constantly because of repeated sequencing of a single patternfor durations ranging from three to over 20 minutes.

One simple solution to overcoming discontinuities in fractional valuesof β>1 is to employ more than one cycle of the lower fundamentalfrequency f_(synth2)=1/T_(synth2) to define the total period of thepattern βT_(synth2). The minimum number of required cycles can bedetermined by converting the decimal ratio into a fraction with thelowest common denominator. This lowest common denominator defines thenumber of cycles of the lower frequency fundamental in the pattern whilethe numerator defines the number of the complete cycles of the higherfrequency.

For example, in the topmost graphic example in FIG. 44 labeledβ=1.5=3/2, two sinusoids having a frequency ratio of 1.5 or fractionallyas 3/2 comprises two-cycles of lower frequency sinusoid f_(synth2) shownby curve 880 and three-cycles of high frequency sinusoid f_(synth1)shown by curve 881 having the same start and end values. Because thecomponent sinusoids start and end with the same value, any chordcombining the two will also be continuous in magnitude and in its slope,i.e. its 1^(st) derivative, across repeated patterns. While the patternmay also be stored comprising an integer multiple of this fraction, e.g.6/4, 12/8, or 24/16, the data sets are substantially larger withoutadding any additional information or improving resolution. Patternscomprising scalar multiples of lowest-common-denominator based fractionsare therefore only beneficial in matching other patterns in a patternlibrary having the same total pattern duration and not for theirfidelity or harmonic content.

Fractions comprising the lowest-common-denominator are applicable forany frequency where the total pattern duration and underlying data fileis manageable. For example, the bottommost graphic example in FIG. 44labeled β=2.33333=7/3 comprises two sinusoids having a frequency ratioof 2.33333 or fractionally as 7/3. In this example, the component of thechords comprise three-cycles of lower frequency sinusoid f_(synth2)shown by curve 882 and seven-cycles of high-frequency sinusoidf_(synth1) shown by curve 883 having the same start and end values.Because the component sinusoids start and end with the same value, anychord combining the two will also be continuous in magnitude and inslope, i.e. in its 1^(st) derivative, across repeated patterns. Becausemore cycles are required to construct a repeating pattern maintainingcontinuity throughout than in the example of where β=1.5, the data fileof such a pattern is naturally larger and longer. While even longduration patterns have manageable file sizes, they are less flexible informing new combinations.

Another means to reduce file size and pattern length is to utilize theprincipal of mirror phase symmetry. For example, in the topmost waveformin FIG. 45 labeled β=11.5 a single-cycle of lower frequency sinusoid ofperiod T_(synth2) is combined with sinusoid 886 having a frequency 11.5times higher. Sinusoid 886 is one-half a cycle short of being 12 fullsinusoidal cycles, as shown by missing piece 887. Even though both sinewaves have the same amplitude at the beginning and end of the pattern,the slope of sinusoid 886 is negative at the end of the pattern, meaningthe function is positive and declining in magnitude at the end of thepattern. Repeating the pattern will result in two positive “humps” inthe sine wave producing an unwanted higher harmonic spectral component.

Rather than doubling the length of the pattern to its lowest commondenominator fraction β=23/2 to avoid this issue, another option is tonumerically synthesize a mirror phase pattern. This inventive method asdisclosed herein is shown in the bottommost graphs of FIG. 45 wherebyfundamental sine wave 885 remains the same in both normal phase andmirror phase patterns, while higher frequency sinusoid 886 shown in thenormal phase pattern is inverted to form sinusoid 888 in the mirrorphase. The alternating combination of normal phase and mirror phasepatterns results in sinusoids continuous in magnitude and in slope, i.e.in its 1^(st) derivative, without the need for storing long inflexiblepatterns in the pattern library.

In the event that frequencies of irregular fractions are combined, itcan be impractical to find a convenient fraction for constructing two ormore sinusoids of full cycles. For example, FIG. 46 illustrates that thefrequency of sinusoid 891 is not an integral or even fractional multipleof the frequency of fundamental sine wave 890. Instead, sinusoid 891exhibits a gap in amplitude 892 between its value at the start of thepattern and the end. Repeating this pattern will result in a severediscontinuity in amplitude and slope at the transition between the endof one pattern and the beginning of the next pattern. Moreover, becauseof the non-integral fractional multiple in frequency β=1.873, even alarge number of cycles will not converge on a discontinuity-freetransition. One brute-force solution is to employ an interpolated gapfill 894 where sinusoid 891 is modified into curve 893 with aconstructed interpolated line segment 895, created manually or by somemathematical means. The shape of interpolated line segment 895 resultsin no discontinuity in the amplitude of the pattern and minimaldiscontinuity in the slope. While the edit does create some harmonics,it can be designed using Fourier analysis to minimize any adverse impactof harmonic spectra.

The disclosed apparatus and methods for synthesizing sinusoidal andchord excitation patterns for LED drive in phototherapy systems usingdigital synthesis were described in the context wherein the referencecurrent used in the LED drive circuitry remained constant throughout thegeneration of various patterns. Changes in frequency, amplitude and DCoffset can all be generated entirely in the digital domain without theuse of analog synthesis. Pure digital synthesis in the context of thisapplication means the use of PWM synthesis not including PCM audiomethods. In contrast, because it employs digital-to-analog conversionoutputting a time-varying analog output, pulse coded modulation isconsidered herein as analog synthesis. Previous sections of thisdisclosure also described a range of options for generating LED driveusing both purely analog and such PCM and other digitized analogsynthesis methods. Nothing in this application precludes the combinationof using both digital and analog synthesis to generate sinusoids andchords thereof.

The discussion of such mixed-mode synthesis is beyond the scope of thisapplication and will not be described further except in the context ofusing the reference current as a means to adjust the full scale value ofsinusoids generated using digital PWM synthesis. An example of thispoint is illustrated in FIG. 47, wherein the topmost waveformillustrates a PWM generated sine wave 902 using pulses 901 of varyingpulse width in accordance with the previously disclosed methods. Asshown, the reference current αI_(ref) has a value 903 a that at time t₁increases to a higher current 903 b. The result of this change inreference current is illustrated in the bottommost graph of FIG. 47showing the LED current resulting from the described synthesiswaveforms.

In the interval prior to time t₁ when the reference current is biased atcurrent 903 a, the full scale output current of the LED driver is shownby line 905 a. After time t₁ when the reference current is increased tocurrent 903 b, the full scale output current of the LED drivercorrespondingly increases to current level 905 b. Since digitalsynthesis only controls the LED enable signal of the driver, the actualcurrent flowing when the LED driver is conducting is set by thereference current value. As a result, prior to time t₁ the peak-to-peakvalue of sinusoid 906 ranges from zero to current level 905 a whileafter time t₁ the peak-to-peak value of sinusoid 907 ranges from zero tocurrent level 905 b, thereby increasing the magnitude of the outputwithout changing the digital pattern code used in sinusoidal synthesis.At the transition at time t₁ a discontinuity 908 may occur, which withcapacitance present in the LED drive circuit may appear filtered intotransition 909. Since changing the reference current is an infrequentevent in phototherapy, the non-repeating transition has no significantimpact on the frequency spectrum of the LED drive.

Bus Architecture Based Control

Referring to FIG. 27A, a distributed LED driver system comprisesseparate digital synthesizers 203 a through 203 n independentlycontrolling the current in multiple channels of LEDs through the enableinput of MOSFET drivers 215 a through 215 n. Constructed using dedicatedcounters and latches, these digital synthesizers can operateindependently but require a proper sequence of PWM codes to berepeatedly loaded into the counters to synthesize the desired sinusoid.In this regard, collectively digital synthesizers 203 therefore requiresome centralized control able to uniquely access each digitalsynthesizer 203 a through 203 n at high speeds. One such means toimplement this kind of control and communication is through a high-speeddigital bus.

As described in the previously-cited U.S. Pat. No. 9,877,361, abus-controlled LED driver is used to generate programmable square wavepulses. By utilizing the methods disclosed herein, any digital pulsedrive circuit used in LED drives may be repurposed for sinusoidalsynthesis. For example the circuit of FIG. 48 illustrates one suchimplementation of an LED driver including a bus-programmable referencecurrent source 930 a comprising a D/A converter 932 a, which converts an8-bit digital word stored in ILED register 931 a into an analog currentαI_(ref) quantized into 256 levels. If greater resolution is required agreater number of bits, e.g. 12 bits for 4096 quantized levels or 16bits for 65,536 quantized levels, may be used.

As shown, the data setting the current αI_(ref) may be loaded into thelatch of ILED register 931 a from a software or firmware programresiding in a central controller or microprocessor 920 and passed toI_(LED) register 931 a through digital communication bus 923. Becausemore than one channel is generally controlled by the samemicrocontroller 920 and connected to the same common data bus 923, adecoder 925 a is included to detect and store “channel-a” only analoginformation into digital registers 931 a (along with digital synthesisdata for registers 927 a and 928 a), thereby ignoring data for otherchannels.

Control of the bus is managed through bus control circuitry 920 bcontained within microcontroller 920. This information is communicatedby a data bus 921 generally using a standardized protocol such as SPI(serial peripheral interface) or other high-speed alternatives to thevarious ICs connected to the bus. Each IC communicates with the busthrough an SPI interface 922 and translates the serial information intoserial or parallel data specifically formatted for communication insidethe integrated circuit, delivering the information to decoder 925 a andother channels through an internal bus 923. Internal bus data structuressuch as internal bus 923 generally comprise parallel data needing alarge number of conductors while system bus protocols such as SPI bus921 used to connect various ICs together generally comprise high-speedserial data in order to minimize the number of connecting wires. Theinformation relayed from microcontroller 920 to SPI interface 922through SPI bus 921, while it could contain algorithmic information andprograms, generally only comprises the operating settings needed toinstruct the LED driver IC how to drive the LEDs, e.g. the register datafor data registers 927 a, 928 a and 930 a. These settings may be storedin tabular form in pattern EPROM 920 a contained within microcontroller920.

In addition to communicating the digital data for ILED register 931 a,the data decoded in decoder 925 a loads on-time data into t_(on)register 927 a and phase delay data into register 928 a Regardless ofhow programmable current control is achieved for each specific channel,the independent control of an array of multiple strings of LEDs can beachieved by combining or integrating multiple channels of the disclosedLED current driver and controlling them from a central controller ormicroprocessor.

For example, microcontroller 920 contains within its pattern library 920a the waveform synthesis algorithms executed by the LED driver channelas shown by precision gate bias and control circuit 935 a andhigh-voltage MOSFET 936 a. This waveform pattern information generatedby microcontroller 920 is relayed from its internal bus controlcircuitry 920 b to one or more LED driver ICs, using the high-speed SPIbus 921. While other digital interfaces may be employed, the SPI bus hasbecome an industry standard in LCD and HDTV backlighting systems, and acommon interface for LED driver ICs in large displays (but not in smalldisplays used in handheld electronics). As such, this drive electronicscan be repurposed for LED drive in phototherapy, and in accordance withthe methods disclosed herein, may be adapted for sinusoidal synthesisdespite the fact that such ICs were never intended for such purposes.

Using the SPI protocol, each LED driver IC has its own unique chip IDcode. All data packets broadcast from microcontroller 920 on SPI bus 921include this unique chip ID in the header of the data stream as an atype of address—an address employed to direct the data to one and onlyone LED driver IC, i.e. the target LED driver IC. Only data matching aparticular chip ID will be processed by the corresponding target LEDdriver IC even though all driver ICs receive the same data broadcast.The chip ID is typically hardware-programmed for each LED driver IC withone or two pins on the IC. Using a four-state input where each pin canbe either grounded, tied to V_(logic), left open, or grounded through aresistor, an multistate analog comparator interprets the analog leveland outputs a 2-bit digital code. Using two pins, a 4-bit binary word(i.e., a binary nibble) uniquely identifies one of 4² or 16 chip IDs.Whenever a data broadcast is received on SPI bus 921 matching the chipID of any specific LED drive, i.e. the specific IC is “selected”,meaning the particular LED driver IC responds to the broadcastinstructions and settings. Data broadcasts whose data header do notmatch a particular LED driver IC's chip ID are ignored. In summary, eachLED driver channel comprising a set of “n” channel drive circuits isgenerally realized as a single integrated circuit with its own unique“chip ID” used to direct instructions from the microcontroller 920directly to that specific IC and to the LED drive channels containedwithin. The same communication from microcontroller 920 is ignored byall other LED drivers made in integrated circuits without the matchingchip ID.

Within a selected LED driver IC, SPI interface 922 receives theinstructions from SPI bus 921 then interprets and distributes thisinformation to decoder 925 a and other channel decoders through internaldigital bus 923, which instructs the individual LED driver channels ondrive conditions (including channel by channel timing and LED biasing).For high-speed data transmission with a minimal number ofinterconnections, internal digital bus 923 comprises some combination ofserial and parallel communication. Since bus 923 is dedicated andinternal to the LED driver of an LED pad, bus 923 may conform to its owndefined standards and is not subject to complying with anypre-established protocol.

The digital information from digital bus 923, once decoded by decoder925 a and other channels, is next passed to digital data registerspresent within each individual LED driver channel. For clarity ofidentification, respective elements within a given channel utilize thesame letter designator as the channel, for example, counter 227 islabeled as 227 a in channel-a and as 227 b in channel-b (not shown).These registers may be realized with S-type or D-type flip-flops, staticlatch circuitry, or SRAM cells known to those skilled in the art.

In the particular driver IC shown, the decoded data for each channelincludes a 12-bit word defining the channel's on-time t_(on), a 12-bitword defining the phase delay ϕ, and a 8-bit word defining the LEDcurrent, stored respectively in t_(on) register 927 a, ϕ register 928 a,and LED register 931 a and corresponding t_(on), ϕ and I_(LED) registersin the other channels (not shown). For example the decoded output ofdecoder 925 a comprising the t_(on), ϕ, and I_(LED) data for channel-ais loaded into registers 927 a, 928 a, and 931 a, respectively.

As previously described, the on-time t_(on) of LED string 940 a, alongwith the signals Clk θ and Sync on clock line 924 combine to set theLEDs' brightness through the corresponding PWM duty factor D, and inwaveform synthesis to set the pulsed frequency f_(synth) of thesynthesized pattern of photoexcitation. While in pulse synthesis thet_(on), ϕ, and I_(LED) data loaded in their corresponding registerschange infrequently, in sinusoidal synthesis they are updated with everySync pulse to load a new PWM value into counter 929 a.

Similarly, the decoded output of decoder 925 b (not shown) comprisingthe t_(on), ϕ, and I_(LED) data for channel-b is loaded into itscorresponding registers 927 b, 928 b, and 931 b (not shown)respectively, and the decoded output of decoder 925 n comprising thet_(on), ϕ, and I_(LED) data for channel-n is loaded into registers 927n, 928 n, and 931 n respectively (also not shown).

These data registers may operate as clocked latches loading data only atpredefined times, e.g. whenever a Sync pulse occurs, or may be changedcontinuously in real-time. Synchronizing the data loading and executionto a clock pulse is known herein as “synchronous” or “latched” operationwhile operating the latches and counter where the data can be changeddynamically at any time is referred to as “asynchronous” or“non-latched” operation. Latched operation limits the maximum operatingfrequency but exhibits greater noise immunity than asynchronousoperation. In this invention disclosure, sinusoidal waveform synthesisperformed by LED drive can be realized by either method—using eitherlatched or asynchronous methods. In display applications, however, onlylatched operation is employed because of an LCD image's severesensitivity to noise.

In non-latched or asynchronous operation, the data received over SPI bus921 for channel-a is decoded and immediately loaded into the t_(on), ϕ,and I_(LED) registers 927 a, 928 a and 931 a and the correspondingregisters in the other channels b through n, up to and includingregisters 927 n, 928 n and 931 n in channel-n. Depending on the LEDdriver IC's implementation, two possible scenarios can occur thereafter.In the first case the count being executed in counter 929 a is allowedto complete its operation, before new data is loaded into counter 927 aand a new count commences.

By example, in non-latched operation data freshly loaded from decoder925 a into t_(on), ϕ, and LED registers 927 a, 928 a, and 931 a wouldwait until the ongoing count in counter 929 a is completed. After thecount is completed the updated data for t_(on) and ϕ in registers 927 aand 928 a are loaded into counter 929 a and simultaneously the updatedLED data in register 931 a is loaded into D/A converter 932 a changingthe bias condition on precision gate bias and control circuit 935 a.After loading the data, counter 929 a commences immediately countingpulses on the Clk θ line of clock line 924, first by turning off LEDstring 940 a if it was on, then counting the number of pulses in ϕregister 928 a before toggling precision gate bias and control circuit935 a and MOSFET 936 a back on. After turning LED string 940 a back on,counter 929 a then counts the number of counts loaded from t_(on)register 927 a on Clk θ line 223 b before shutting LED string 940 a offagain. The counter 929 a then waits for another instruction.

In the second alternative for non-latched or asynchronous operation thesystem behaves exactly the same as the non-latched operation describedpreviously except that whenever an instruction is received via abroadcast on SPI bus 921, the latch is immediately rewritten andsimultaneously restarted. Other than cutting short the ongoing countcycle at the time the register data was rewritten, the operatingsequence is identical. Regardless of which asynchronous method is used,it takes time to broadcast, decode, and commence operation for each andevery channel on a one-by-one basis. In display applications, the delayin writing new data (and changing an LED string's operating conditions)between the first and last channel of an LCD panel may result in flickerand jitter. As such, asynchronous operation is not a viable option inLCD backlighting. In LED phototherapy, however, where a fixed conditionmay be maintained for minutes, non-latched operation is a viable optionespecially for generating higher frequency LED excitation patterns, i.e.for higher values of f_(synth).

Unlike in asynchronous operation, where data is updated continually, inlatched or synchronous operation the LED operating conditions areupdated only a predetermined occasions, either synchronized to fixedtimes, or prescribed events. In latched operation of the circuit shownin FIG. 48, whenever the Sync pulse occurs on line 924, the data mostrecently loaded into t_(on) register 927 a and ϕ register 928 a isloaded into counter 929 a. Counter 929 a then commences counting anumber of pulses on the Clk θ line 924 equal to the number stored in ϕregister 928 b before toggling precision gate bias and control circuit935 a on. After completing the count, the counter 927 a toggles onprecision gate bias and control circuit 935 a, biasing the gate ofcurrent sink MOSFET 936 a to conduct a prescribed amount of currentI_(LEDa) thereby illuminating LED string 940 a to a desired level ofbrightness. Counter 929 a subsequently counts the number of Clk θ pulsesloaded from t_(on) register 927 a until the count is complete, and thentoggles precision gate bias and control circuit 935 a to shut offcurrent MOSFET 936 a and terminate illumination. At this point,depending on LED driver IC's design, LED string 940 a may remain off forthe remainder of the T_(sync) period, i.e. until the next Sync pulseappears on clock line 924, or alternatively repeatedly toggle on and offat the value loaded into t_(on) register 927 a until the next Sync pulseoccurs on line 223 a.

In latched systems the Sync pulse serves several purposes. First, it isan instruction to load the data from the t_(on) register 927 a and the ϕregister 928 a into the programmable digital counter 227 a. Second, itis an instruction to reset the counter 929 a and commence counting incounter 929 a, first to pass a period of time corresponding to the phasedelay ϕ, and then to turn on the LED string 940 a for the number ofclock counts loaded into the corresponding t_(on) register 927 a.Thirdly, it is an instruction to load the value in the I_(LED) register931 a into the D/A converter 932 a, precisely setting the analog valueof current αL_(ref). Similar operations are performed in thecorresponding counters, D/A converters, and t_(on), ϕ and I_(LED)registers in the other channels. Finally, it prevents noise fromoverwriting the data in the registers 927 a, 928 a and 931 a midstreamjumbling the count.

Phototherapeutic Strategy

Using the described inventions to facilitate sinusoidal synthesis of LEDdrive and illumination patterns for phototherapy applications,photobiological processes in tissue repair and immune response can bestimulated with a greater degree of precision, control and tissuespecificity, free from spectral contamination present in pulsed LEDdrives. The generation of sinusoidal drive waveforms may be performedusing analog synthesis, digitally-controlled analog synthesis (PCM), orby purely digital synthesis methods, preferably using fixed frequencyPWM techniques. The LED driving waveforms may include a simultaneous mixand/or a programmed sequence of audio-frequency square wave pulses, sinewaves, chords of sinusoids, and any other time-varying waveforms such asramp and triangle waves, filtered audio sources, or combinationsthereof.

The disclosed methods may be used for driving any wavelength LED orlaser diode, including long infrared, near infrared, visible lightincluding deep red, red, blue and violet, as well as driving nearultra-violet LEDs. Far UV and beyond are excluded because of thedetrimental health risks of ionizing radiation.

As disclosed, the methods and apparatus facilitate control of keyparameters for phototherapy, namely

-   -   Magnitude of oscillating LED current drive (AC amplitude)    -   Frequencies of synthesized sinusoidal oscillations in LED drive    -   Magnitude of continuous LED current drive (DC offset)    -   Chords of multiple sinusoidal frequencies        The control may be performed dynamically or in prescribed        patterns made in advance of their use and stored in pattern        libraries. By controlling the above variables without the        potential adverse impact of unwanted audio frequency harmonics,        particularly of odd harmonic multiples, a strategy consistent        with the principles of bioresonance and photobiological time        constants can be realized.

An example of a phototherapeutic strategy is graphically illustrated in3D in FIG. 49, where the x-axis represents the peak-to-peak amplitude ofan oscillating LED current from 0 mA to 30 mA, the y-axis represents theconstant DC component of the LED current ranging from 0 mA to 30 mA, andthe z-axis represents the AC frequency of sinusoidal oscillationsranging from 0.1 Hz (nearly DC) to over 10 kHz. The locations of thevarious physiological structures and conditions, shown by the numerals960 through 983, illustrate the areas of possible maximum beneficialeffects from particular combinations of the amplitude, sinusoidalfrequency and DC component of the current used to illuminate the LEDstring. The graph illustrates in general terms the prior observationthat electron transport 960 can occur at higher frequencies, in therange of kHz and beyond, ionic transport 961 occurs in tens-to-hundredsof Hertz, and chemical transformations 962 occur in the single-digitHertz range. Also in the single-digit range, albeit specifically athigher DC currents or higher low-frequency AC currents, transientthermal effects are manifest. Steady state thermal processes 964 occurat even higher DC currents from increased heating at frequencies from0.1 Hz to DC, i.e. 0 Hz.

Also, as shown, higher magnitude AC is required to stimulate entireorgans 967, while lesser current is need to treat patches of tissue 966,and even smaller current to affect concentrated groups of cells 965.Using too high of AC amplitude may actually reduce efficacy byintroducing energy at a rate higher than a specific photobiologicalprocess can absorb or use. Among the treatments shown in an exemplaryfashion in FIG. 49, muscles 970 and thermotherapy 969 benefit fromgreater heating and therefore require a higher continuous LEDillumination, i.e. a greater DC offset.

Neurological response such as neural 982 and relaxation 981 benefitsfrom higher frequencies and moderate AC currents with minimal DC offset.Photodynamic therapy 980, where photons are being used to stimulate oractivate a photochemical process, or anti-bacterial treatments whereenergy is attempting to impede normal bacterial metabolism require acombination of high excitation frequencies and high AC LED current.Photodynamic therapy also benefits from high total light intensity,meaning brighter and hence higher DC currents are better.

At moderate frequencies and AC current levels with little or no DCcontents a variety of remedies exist including treatment therapies forcirculation and angiogenesis 974, immune system and hormonal stimulation973 and skin 972, exhibiting treatment mechanisms at both the cellularand tissue level. Lungs 971, heart, kidney, liver, pancreas and othermajor bodily organs benefit from an increased AC current invokingmechanisms at both the tissue and organ levels.

Regardless of whether the specific treatments offer efficaciesconsistent with the 3D graph as depicted, prior pulsed light experimentswith their spectral contamination still reveal a significant influenceof pulse frequency and LED brightness of treatment efficacy. Using theanalog and digital synthesis methods disclosed herein, the ability ofthe disclosed apparatus of methods to generate and control the frequencyand amplitude of sinusoidal excitation of LEDs is expected to profoundlyimprove phototherapy control and efficacy beyond that of any prior artdigitally pulsed LED or laser system.

We claim:
 1. A phototherapy process comprising: providing a flexible LEDpad, the flexible LED pad comprising a plurality of light-emittingdiodes (LEDs); positioning the flexible LED pad adjacent the skin of aliving human being or animal; causing the LEDs to emit light through theskin into the human being or animal so as to produce a medicallytherapeutic effect in an organ, tissue or physiological system of thehuman being or animal by means of a photobiomodulation process; andvarying an intensity of the light emitted by the LEDs in accordance witha sinusoidal function, wherein the sinusoidal function comprises a chordcomprising a plurality of sine waves, each of the sine waves in thechord having a frequency in the audio range, the frequency of each ofthe sine waves the chord being different from the frequency of each ofthe other sine waves in the chord.
 2. The phototherapy process of claim1 wherein the frequency of each of the sine waves is greater that 20 Hzand less than 20 kHz.
 3. The phototherapy process of claim 1 whereinvarying an intensity of light emitted by the LEDs in accordance with asinusoidal function comprises connecting the LEDs to a controlledcurrent element, the controlled current element comprising a currentsource or a current sink, the controlled current element operatingthrough feedback to maintain a current of a prescribed magnitude in theLEDs.
 4. The phototherapy process of claim 1 comprising creating thechord by delivering an electrical signal representing each of theplurality of sine waves to an analog mixer.
 5. The phototherapy processof claim 1 comprising varying the frequency of at least one of the sinewaves while the process is being performed on the human being or animal.6. The phototherapy process of claim 1 comprising creating the chord bystrobing an analog sine waveform ON and OFF at a strobe frequency. 7.The phototherapy process of claim 1 wherein varying an intensity oflight emitted by the LEDs in accordance with a sinusoidal functioncomprises: connecting the LEDs to a controlled current element, thecontrolled current element comprising a current source or a currentsink, the controlled current element operating through feedback tomaintain a current of a prescribed magnitude in the LEDs when thecontrolled current element is turned on and to prevent current fromflowing in the LEDs when the controlled current element is turned off;and controlling the controlled current element such that the LEDs emitlight in accordance with the sinusoidal function.
 8. The phototherapyprocess of claim 7 wherein controlling the controlled current elementcomprises: generating a reference current; and using the controlledcurrent element to provide a current in the LEDs having a magnitudegreater than a magnitude of the reference current by a predeterminedratio.
 9. The phototherapy process of claim 8 wherein the controlledcurrent element comprises a first current mirror MOSFET and a secondcurrent mirror MOSFET, the first current mirror MOSFET being thresholdconnected, the process comprising: causing the reference current to flowthrough the first current mirror MOSFET; and causing the current in theLEDs to flow through the second current mirror MOSFET.
 10. Thephototherapy process of claim 9 wherein the controlled current elementcomprises a current control MOSFET, the process comprising causing thecurrent in the LEDs to flow through the current control MOSFET.
 11. Thephototherapy process of claim 10 wherein varying an intensity of lightemitted by the LEDs in accordance with a sinusoidal function comprisesusing the current control MOSFET to switch the current in the LEDs ONand OFF so as to generate a sequence of current pulses, each pulse inthe sequence having a duty factor equal to t_(on)/T_(sync), whereinT_(sync) represents a time between a leading edge of the pulse and aleading edge of a next pulse in the sequence and t_(on) represents aduration of the pulse, the respective duty factors of the pulsesrepresenting values of the sinusoidal function at successive points intime.
 12. The phototherapy process of claim 11 wherein a sync frequencyis greater than 20 kHz, the sync frequency being equal to 1/T_(sync).13. The phototherapy process of claim 11 wherein switching the currentin the LEDs ON and OFF so as to generate a sequence of current pulsescomprises applying an enable signal to the gate terminal of the currentcontrol MOSFET so as to turn the current control MOSFET ON and OFF. 14.The phototherapy process of claim 13 comprising generating the enablesignal as a series of pulses, the leading edges of the pulses of theenable signal being separated by T_(sync), each of the pulses of theenable signal having a duration equal to T_(on).
 15. The phototherapyprocess of claim 14 comprising varying the reference current inaccordance with the sinusoidal function and generating the pulses of theenable signal from the reference current such that the frequency of theenable signal is an integral multiple of the frequency of the sinusoidalfunction.
 16. The phototherapy process of claim 14 wherein applying anenable signal to the gate terminal of the current control MOSFETcomprises: loading data representing T_(sync) into a T_(sync) counter;loading data representing T_(on) into a pulse-width modulation (PWM)counter; supplying clock pulses at a clock frequency f_(θ) to theT_(sync) and PWM counters; turning the current control MOSFET ON;decrementing the T_(sync) and PWM counters at the clock frequency f_(θ)when the current control MOSFET is turned on; and turning the currentcontrol MOSFET OFF when the data in the PWM counter reaches a firstpreselected value.
 17. The phototherapy process of claim 16 comprising,after turning the current control MOSFET OFF, turning the currentcontrol MOSFET ON again when the data in the T_(sync) counter reaches asecond preselected value.
 18. The phototherapy process claim 17 whereineach of the first and second preselected values is equal zero.
 19. Thephototherapy process of claim 16 wherein the clock frequency f_(θ) isgreater than 20 kHz.
 20. The phototherapy process of claim 10comprising: detecting a voltage difference between a first voltage at adrain terminal of the first current mirror MOSFET and a second voltageat a drain terminal of the second current mirror MOSFET; using thevoltage difference to generate a gate drive voltage; and delivering thegate drive voltage to a gate terminal of the current control MOSFET. 21.The phototherapy process of claim 20 comprising varying the referencecurrent in accordance with the sinusoidal function.
 22. The phototherapyprocess of claim 21 wherein varying the reference current in accordancewith the sinusoidal function comprises supplying an input terminal of adigital-to-analog (D/A) converter with a series of digital values, eachof the digital values representing a value of a sine wave at an instantin time.
 23. The phototherapy process of claim 9 wherein thepredetermined ratio is equal to a ratio of the respective gate widths ofthe first and second current mirror MOSFETs.
 24. The phototherapyprocess of claim 1 comprising generating each of the sine waves usingpulse width modulation with pulses at a frequency f_(sync), wherein afrequency of each of the plurality of sine waves is less than 20 kHz andthe frequency f_(sync) is greater than 20 kHz.
 25. The phototherapyprocess of claim 1 wherein at least one of the plurality of sine waveshas an audio frequency selected from the group consisting of 292 Hz andintegral multiples of 292 Hz.
 26. The phototherapy process of claim 1wherein the photobiomodulation process produces a photobiologicalprocess within a mitochondrion in a eukaryotic cell in the human beingor animal.
 27. The phototherapy process of claim 26 wherein thephotobiomodulation process comprises a photon impinging on acytochrome-c oxidase (CCO) molecule within a mitochondria of a cell,thereby increasing the energy content of the cell by transforming anadenosine monophosphate (AMP) molecule into an adenosine diphosphate(ADP) molecule, the ADP molecule having an energy higher than an energyof the AMP molecule, and converting the ADP molecule into an adenosinetriphosphate (ATP) molecule, the ATP molecule having an energy higherthan the energy of the ADP molecule.
 28. The phototherapy process ofclaim 27 wherein the photobiomodulation process comprises the release ofnitric oxide from the CCO molecule.